Integrated VCO having an improved tuning range over process and temperature variations

ABSTRACT

An integrated VCO having an improved tuning range over process and temperature variations. There is therefore provided in a present embodiment of the invention an integrated VCO. The VCO comprises, a substrate, a VCO tuning control circuit responsive to a VCO state variable that is disposed upon the substrate, and a VCO disposed upon the substrate, having a tuning control voltage input falling within a VCO tuning range for adjusting a VCO frequency output, and having its tuning range adjusted by the tuning control circuit in response to the VCO state variable.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of Ser. No. 09/580,014, filed on May26, 2000, now U.S. Pat. No. 6,426,680 which claims the benefit of U.S.Provisional Application No. 60/136,116 filed on May 26, 1999, thecontents of both of which are hereby incorporated by reference.

The Ser. No. 09/580,014 application is a continuation-in-part of patentapplication entitled “System and Method for Linearizing a CMOSDifferential Pair,” by Haideh Khorramabadi, filed May 17, 2000, U.S.patent application Ser. No. 09/573,356, which is a continuation-in-partof application Ser. No. 09/547,968, filed Apr. 12, 2000; which is acontinuation-in-part of application Ser. No. 09/493,942, filed Jan. 28,2000; which is a continuation-in-part of application Ser. No.09/483,551, filed Jan. 14, 2000; which is a continuation-in-part ofapplication Ser. No. 09/439,101 filed Nov. 12, 1999; the disclosures ofwhich are incorporated herein by reference.

BACKGROUND OF THE INVENTION

Voltage control oscillators (VCO) play a key role in a number ofapplications which require a variable frequency source, such as phaselocked loops, and the like. In an exemplary phase lock loop (PLL), a VCOmight be used as an oscillator that produces a variable frequency ascontrolled by a filtered output from a phase detector. A phase lock loopis a device that is capable of being programmed to generate a desiredfrequency. Once it approximates the frequency, the frequency is divideddown to a value of a reference frequency, provided by an externaloscillator, and compared to that reference frequency. When thedifference between the reference frequency and the generated frequencyreaches zero, the phase lock loop stops tuning and locks to thefrequency that it has just produced. The frequency reference used totune the phase lock loop is typically provided by a single frequencyoscillator circuit. Alternatively, a series of multiple oscillators maybe used to provide a range of frequencies. However, the multipleoscillator approach would not provide a continuously variable frequencyoutput.

A VCO, typically found in a PLL, generates signals that are phasedlocked to a reference frequency. Thus, the VCO must provide an outputsignal phase locked to the reference frequency.

VCOs contained in PLLs are also required to produce a stable referencefrequency. A low phase noise present in the VCO output is desirable toproduce a stable output.

VCOs typically accomplish a change in frequency by changing a value ofcapacitance in the VCO circuitry. Capacitors may be discretely switchedin by mechanical switches, or may be a continuously variable mechanicaltype, or as is typically the case, a variable capacitance is provided bya device that provides a capacitance proportional to an applied voltage.In a given circuit, variations and component values will require anadjustable capacitance that will compensate for a component spread toprovide a variable capacitance that allows a desired range of tuningfrequencies to be produced by the VCO. The effects of component spreadare magnified when a VCO is disposed on an integrated circuit substrate.Components constructed on an integrated circuit substrate typically havewide variations and component values. Thus, the variable capacitancedevices used to tune voltage control oscillators are required to have awider tuning range of capacitances to compensate for the spread inintegrated circuit component values.

Those having skill in the art would understand the desirability ofhaving a voltage control oscillator capable of being integrated on anintegrated circuit substrate that does not have the problem of requiringa large variable capacitance tuning range. This type of device wouldnecessarily provide higher integrated circuit yields by providing a VCOrequiring less of a variable tuning voltage range, thus allowing atunable VCO to be economically integrated onto an integrated circuit.

SUMMARY OF THE INVENTION

There is therefore provided in a present embodiment of the invention anintegrated VCO. The VCO comprises, a substrate, a VCO tuning controlcircuit responsive to a VCO state variable that is disposed upon thesubstrate, and a VCO disposed upon the substrate, having a tuningcontrol voltage input falling within a VCO tuning range for adjusting aVCO frequency output, and having its tuning range adjusted by the tuningcontrol circuit in response to the VCO state variable.

Many of the attendant features of this invention will be more readilyappreciated as the same becomes better understood by reference to thefollowing detailed description considered in connection with theaccompanying drawings.

DESCRIPTION OF THE DRAWINGS

These and other features and advantages of the present invention will bebetter understood from the following detailed description read in lightof the accompanying drawings, wherein

FIG. 1 is an illustration of a portion of the over-the-air broadcastspectrum allocations in the United States;

FIG. 2 is an illustration of the frequency spectrum of harmonicdistortion products;

FIG. 3 is an illustration of a spectrum of even and odd orderintermodulation distortion products;

FIG. 4 is an illustration of interference caused at the IF frequency bya signal present at the image frequency;

FIG. 5 is an illustration of a typical dual conversion receiverutilizing an up conversion and a subsequent down conversion;

OSCILLATOR FIGURES

FIG. 6 is a semi-schematic simplified timing diagram of differentialsignals, including a common mode component, as might be developed by adifferential crystal oscillator in accordance with the invention;

FIG. 7 is a semi-schematic block diagram of a differential crystaloscillator, including a quartz crystal resonator and oscillator circuitdifferentially coupled to a linear buffer amplifier in accordance withthe invention;

FIG. 8 is a simplified schematic illustration of differential signalspresent at the output of a crystal resonator;

FIG. 9 is a simplified schematic diagram of a quartz crystal resonatorequivalent circuit;

FIG. 10 is a simplified graphical representation of a plot of impedancevs. frequency for a crystal resonator operating near resonance;

FIG. 11 is a simplified graphical representation of a plot of phase vs.frequency for a crystal resonator operating near resonance;

FIG. 12 is a simplified schematic diagram of the differential oscillatorcircuit of FIG. 7;

FIG. 13 is a simplified, semi-schematic block diagram of a periodicsignal generation circuit including a crystal oscillator having balanceddifferential outputs driving cascaded linear and non-linear bufferstages;

FIG. 14 is a simplified schematic diagram of a differential foldedcascade linear amplifier suitable for use in connection with the presentinvention;

FIG. 15 is a simplified, semi-schematic diagram of a differentialnonlinear buffer amplifier suitable for use as a clock buffer inaccordance with the invention;

FIG. 16 is a semi-schematic illustration of an alternative embodiment ofthe differential oscillator driver circuit;

FIG. 17 is an block diagram of a differential crystal oscillator as areference signal generator in a phase-lock-loop;

FIG. 18 is a simplified block diagram of an illustrative frequencysynthesizer that might incorporate the differential periodic signalgeneration circuit of the invention;

COARSE/FINE PLL TUNING FIGURES

FIG. 19 is a block diagram illustrating the exemplary frequencyconversions for receiver tuning utilized in the embodiments of theinvention;

FIG. 20 is a block diagram of an exemplary tuner designed to receive a50 to 860 MHz bandwidth containing a multiplicity of channels;

FIG. 21 is an exemplary table of frequencies utilizing coarse and finePLL tuning to derive a 44 MHz IF;

FIG. 22 is an illustration of an alternative embodiment of the coarseand fine PLL tuning method to produce an exemplary final IF of 36 MHz;

FIG. 23 is a block diagram of a dummy component used to model anoperative component on an integrated circuit chip;

FILTER TUNING FIGURES

FIG. 24a is a block diagram of a tuning process;

FIG. 24b is a flow diagram of the tuning process;

FIG. 24c is an exemplary illustration of the tuning process;

FIG. 25 is a block diagram of an exemplary tuning circuit;

FIG. 26 illustrates the amplitude and phase relationship in an LC filterat resonance;

FIG. 27 is a schematic diagram showing the configuration of switchablecapacitors in a differential signal transmission embodiment;

ACTIVE FILTER MULTI-TRACK INTEGRATED SPIRAL INDUCTOR FIGURES

FIG. 28a is a plan view of a multi-track spiral inductor suitable forintegration onto an integrated circuit, such as one produced with a CMOSprocess;

FIGS. 28b-28 g illustrate various planar devices comprising inductor andtransformer configurations suitable for incorporating multiple tracksinto their designs;

FIG. 28h is an illustration of a second embodiment of an inductor havinga single winding comprising five tracks per layer;

FIG. 28i illustrates the placement of tracks in a layered structure;

FIG. 28j is an illustration of an embodiment utilizing a shield disposedbeneath an inductor;

FIG. 28k is an illustration of a patterned shield 2864 that is utilizedbeneath a multi-track inductor;

FIG. 29 is an illustration of the effect of decreasing “Q” on theselectivity of a tuned circuit;

FIG. 30 is an illustration of a typical filter bank utilized inembodiments of the invention for filtering I and Q IF signals;

ACTIVE FILTER UTILIZING A LINEARIZED DIFFERENTIAL PAIR AMPLIFIER FIGURES

FIG. 31a is a diagram of an exemplary differential transconductancestage with an LC load;

FIG. 31b is a block diagram of a linearized differential pair amplifierthat is coupled to distortion canceling linearization circuit;

FIG. 31c is an illustration depicting a representative channel of anyone of the typical field effect of transistors M1 M2, M3, M4;

FIG. 31d is a block diagram showing the interconnection of adifferential pair amplifier to a linearization circuit;

FIG. 31e is a schematic illustrating a CMOS differential pair oftransistors;

FIG. 31f is a graph of a differential current (ΔI_(1,2)=ΔI_(d)) andnormalized transconductance (G_(m)/g_(m)) as input voltage (V_(in=ΔV)_(i)) is varied in the differential pair of FIG. 31e;

FIG. 31g is a schematic diagram of a differential pair amplifier 3127with a second cross coupled differential pair error amplifier added thattends to reduce distortion;

FIG. 31h is a graph illustrating The linearized output current of across coupled differential output amplifier;

FIG. 31i is a schematic of a differential pair amplifier incorporatingtwo auxiliary cross-coupled differential pairs to improve linearizationof the output response I₁ and I₂;

FIG. 31j is a graph of the currents present in the main and twoauxiliary differential pair amplifiers graphed against input voltage asmeasured across the input terminals where Vin=V_(i1)−V_(i2);

FIG. 31k is a graph of transconductance curves for the differentialamplifier made up of a main differential pair amplifier 3103 and alinearization circuit comprising differential pair amplifiers;

FIG. 31l illustrates an equivalent circuit that provides an offsetvoltage V_(os) that permits shaping of The G_(m) ^(Total) curve;

FIG. 31m is a graph of the transconductance curve for The exemplarydifferential pair amplifier that extends the input voltage range byallowing ripple in the overall G_(m) of the amplifier;

FIG. 32 shows a transconductance stage with an LC load and Qenhancement;

ACTIVE FILTER INDUCTOR Q TEMPERATURE COMPENSATION FIGURE

FIG. 33 shows a method of tuning inductor Q over temperature;

COMMUNICATIONS RECEIVER FIGURE

FIG. 34 is a block diagram of a communications network utilizing areceiver according to any one of the exemplary embodiments of theinvention;

RECEIVER FRONT END-PROGRAMABLE ATTENUATOR AND LNA FIGURES

FIG. 35 is an is an illustration of the input and output signals of theintegrated switchless programmable attenuator and low noise amplifier;

FIG. 36 is a functional block diagram of the integrated switchlessprogrammable attenuator and low noise amplifier circuit;

FIG. 37 is a simplified diagram showing the connection of multipleattenuator sections to the output of the integrated switchlessprogrammable attenuator and low noise amplifier;

FIG. 38 is an illustration of an exemplary embodiment showing how theattenuator can be removed from the circuit so that only the LNAs areconnected;

FIG. 39 is an attenuator circuit used to achieve one dB per stepattenuation;

FIG. 40 is an exemplary embodiment of an attenuator for achieving afiner resolution in attenuation then shown in FIG. 5;

FIG. 41 is an illustration of the construction of series and parallelresistors used in the attenuator circuit of the integrated switchlessprogrammable attenuator and low noise amplifier;

FIG. 42 is an illustration of a preferred embodiment utilized to turn oncurrent tails of the differential amplifiers;

FIG. 43 is an illustration of an embodiment showing how the individualcontrol signals used to turn on individual differential pair amplifiersare generated from a single control signal;

FIGS. 44a and 44 b are illustrations of an embodiment of comparatorcircuitry used to activate individual LNA amplifier stages;

RECEIVER FREQUENCY PLAN AND FREQUENCY CONVERSION LOCAL OSCILLATORRELATIONSHIP FIGURE

FIG. 45a is a block diagram illustrating the exemplary generation of thelocal oscillator signals utilized in the embodiments of the invention;

NARROW BAND PLL2 AND VCO FIGURES

FIG. 45b is a block diagram that illustrates the relation of the VCO tothe second LO generation by PLL2.

FIG. 45c is a block diagram of an embodiment of a VCO utilizing a tuningcontrol circuit;

FIG. 45d is a block diagram of an embodiment of a VCO utilizing a tuningcontrol circuit showing tuning control circuit interaction with majorVCO components;

FIG. 45e is a schematic of a feedback network that allows the frequencyof oscillation to be adjusted;

FIG. 45f is a schematic of a feedback network that allows the frequencyof oscillation to be adjusted by varactor tuning including NMOS devices;

FIG. 45g is a graph of capacitance verses control voltage applied to anNMOS varactor;

FIG. 45h is a graph illustrating average capacitance achievable with anNMOS varactor;

FIG. 45i is a schematic of an embodiment of a VCO;

FIG. 45j is a schematic of an equivalent circuit of the VCO of FIG. 45i;

FIG. 45k is a schematic of a tuning control circuit controlling switchedcapacitors to center a varactor tuning range;

FIG. 46a is a schematic of a PLL having its VCO controlled by anembodiment of a VCO tuning control circuit;

FIG. 46b illustrates a pulse train output of the phase detector;

NARROW BAND VCO TUNING FIGURES

FIG. 47a is a process flow diagram illustrating the process of tuningthe VCO with an embodiment of a VCO control circuit;

FIG. 47b is a flow diagram of a PLL start up and locking process for anembodiment of the invention;

FIG. 47c is a graph of a family of frequency verses control voltage forvarious capacitor values that illustrates the use of comparatorhysteresis to aid in achieving a frequency lock condition;

FIG. 47d is a graph of a family of frequency verses control voltage forvarious capacitor values that illustrates the use of dual comparatorwindows to aid in achieving a frequency lock condition;

RECEIVER FIGURES

FIG. 48 is a block diagram of the first exemplary embodiment of theinvention;

FIG. 49 is an illustration of the frequency planning utilized in theexemplary embodiments of the invention;

FIG. 50 is a block diagram showing how image frequency cancellation isachieved in an I/Q mixer;

FIG. 51 is a block diagram of the second exemplary embodiment of thepresent invention;

FIG. 52 is a block diagram of the third exemplary embodiment of thepresent invention;

FIG. 53 is a block diagram of a CATV tuner that incorporates the fullyintegrated tuner architecture;

TELEPHONY OVER CABLE EMBODIMENT FIGURE

FIG. 54 is a block diagram of a low power embodiment of the receiverthat has been configured to receive cable telephony signals.

ELECTRONIC CIRCUITS INCORPORATING EMBODIMENTS OF THE RECEIVER FIGURES

FIG. 55 is a block diagram of a set top box that incorporates thereceiver embodiments;

FIG. 56 is a block diagram of a television that incorporates thereceiver embodiments;

FIG. 57 is a block diagram of a VCR that incorporates the receiverembodiments;

FIG. 58 is a block diagram of a cable modem that incorporates theintegrated switchless programmable attenuator and low noise amplifier;

ESD PROTECTION FIGURES

FIG. 59 is an illustration of a typical integrated circuit die layout;

FIG. 60 illustrates an embodiment of the invention that utilizes padring power and ground busses;

FIG. 61 is an illustration of the connection of a series of powerdomains to a pad ring bus structure;

FIG. 62 is an illustration of an embodiment utilizing an ESD groundring;

FIG. 63 is an illustration of the effect of parasitic circuit elementson an RF input signal;

FIG. 64 illustrates a cross-talk coupling mechanism;

FIG. 65 is an illustration of an ESD device disposed between aconnection to a bonding pad and power supply traces;

FIG. 66 is an illustration of parasitic capacitance in a typical bondingpad arrangement on an integrated circuit;

FIG. 67 is an illustration of a embodiment of a bonding pad arrangementtending to reduce parasitic capacitances;

FIG. 68 illustrates a cross section of the bonding pad structure of FIG.67;

FIGS. 69a-69 e illustrate various ESD protection schemes utilized in thestate of the art to protect an integrated circuit from ESD discharge dueto charge build up on a die pad;

FIG. 70 illustrates an approach to pad protection during ESD event;

FIG. 71 is a schematic of a circuit immune to noise that uses an ggNMOS′C_(gd) and a gate boosting structure to trigger ESD protection;

FIG. 72 is a schematic of an alternative embodiment utilizing the gateboosting structure and a cascode configuration; and

FIG. 73 is a schematic of an embodiment that does not require a quietpower supply.

IF AGC AMPLIFIER FIGURES

FIG. 74 is a block diagram of a variable gain amplifier (“VGA”);

FIG. 75, is a block diagram of the internal configuration of the VGA andthe linearization circuit;

FIG. 76 is a graph of gain versus the control current iSig. Controlcurrent iSig is shown as a fraction of iAtten, with the total currentbeing equal to 1, or 100%;

FIG. 77 is the schematic diagram of an embodiment of the VGA. The VGAhas a control circuit to control the V_(ds) of M10 and M13 at node 7505,and the V_(ds) of M4 and B14 at node 7507;

FIG. 78a illustrates a family of curves showing the relationship of atransistor's drain current (“I_(d)”) to its gate source voltage(“V_(gs)”) measured at each of a series of drain source voltages(“V_(ds)”) from 50 mV to 1 V;

FIG. 78b is a graph of g_(m) verses V_(gs) as V_(ds) is varied from 50mV to 1 V;

FIG. 78c is a graph of the cross-section of FIG. 78b plotting g_(m)verses V_(ds) for various values of V_(gs);

FIG. 79 is a schematic of a current steering circuit; and

FIG. 80 is a schematic of a VD1 control signal generation circuit.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 is an illustration of a portion of the radio frequency spectrumallocations by the FCC. Transmission over a given media occurs at anyone of a given range of frequencies that are suitable for transmissionthrough a medium. A set of frequencies available for transmission over amedium are divided into frequency bands 102. Frequency bands aretypically allocations of frequencies for certain types of transmission.For example FM radio broadcasts, FM being a type of modulation, isbroadcast on the band of frequencies from 88 MHz to 108 MHz 104.Amplitude modulation (AM), another type of modulation, is allocated thefrequency band of 540 kHz to 1,600 kHz 106. The frequency band for atype of transmission is typically subdivided into a number of channels.A channel 112 is a convenient way to refer to a range of frequenciesallocated to a single broadcast station. A station broadcasting on agiven channel may transmit one or more radio frequency (RF) signalswithin this band to convey the information of a broadcast. Thus, severalfrequencies transmitting within a given band may be used to conveyinformation from a transmitter to a broadcast receiver. For example, atelevision broadcast channel broadcasts its audio signal(s) 108 on afrequency modulated (FM) carrier signal within the given channel. A TVpicture (P) 110 is a separate signal broadcast using a type of amplitudemodulation (AM) called vestigial side band modulation (VSB), and istransmitted within this channel.

In FIG. 1 channel allocations for a television broadcast band showingthe locations of a picture and a sound carrier frequencies within achannel are shown. Each channel 112 for television has an allocatedfixed bandwidth of 6 MHz. The picture 110 and sound 108 carriers areassigned a fixed position relative to each other within the 6 MHz band.This positioning is not a random selection. The picture and soundcarriers each require a predetermined range of frequencies, or abandwidth (BW) to sufficiently transmit the desired information. Thus, achannel width is a fixed 6 MHz, with the picture and sound carrierposition fixed within that 6 MHz band, and each carrier is allocated acertain bandwidth to transmit its signal.

In FIG. 1 it is seen that there are gaps between channels 114, and alsobetween carrier signals 116. It is necessary to leave gaps of unusedfrequencies between the carriers and between the channels to preventinterference between channels and between carriers within a givenchannel. This interference primarily arises in the receiver circuit thatis used to receive these radio frequency signals, convert them to ausable frequency, and subsequently demodulate them.

Providing a signal spacing allows the practical design andimplementation of a receiver without placing unrealistic requirements onthe components in the receiver. The spaces help prevent fluctuations inthe transmission frequency or spurious responses that are unwantedbyproducts of the transmission not to cause interference and signaldegradation within the receiver. Also, signal spacing allows the designrequirements of frequency selective circuits in the receiver to berelaxed, so that the receiver may be built economically while stillproviding satisfactory performance. These spectrum allocations andspacings were primarily formulated when the state of the art in receiverdesign consisted of discrete components spaced relatively far apart on aprinted circuit board. The increasing trend towards miniaturization haschallenged these earlier assumptions. The state of the art in integratedcircuit receiver design has advanced such that satisfactory performancemust be achieved in light of the existing spectrum allocations andcircuit component crowding on the integrated circuit. New ways ofapplying existing technology, as well as new technology are continuallybeing applied to realize a miniaturized integrated receiver thatprovides satisfactory performance. Selectivity is a principal measure ofreceiver performance. Designing for sufficient selectivity not onlyinvolves rejecting other channels, but the rejection of distortionproducts that are created in the receiver or are part of the receivedsignal. Design for minimization or elimination of spurious responses isa major objective in state of the art receiver design.

FIG. 2 is an illustration of harmonic distortion products. Transmittedspurious signals, and spurious signals generated in a receiver, mostcommonly consist of harmonics created by one frequency andintermodulation distortion, created by the interaction of multiplefrequencies. Spurious signals at other than the desired frequency arisefrom the inherent nonlinear properties in the circuit components used.These nonlinearities can not be eliminated, but by careful engineeringthe circuitry can be designed to operate in a substantially linearfashion.

When a single frequency called a fundamental 202 is generated, unwantedspurious signals 204 are always generated with this fundamental. Thespurious signals produced as a result of generating a single frequency(f) 202 are called harmonics 204 and occur at integer multiples of thefundamental frequency (2f, 3f, . . . ) The signal strength or amplitudeof these harmonics decrease with increasing harmonic frequency.Fortunately these distortion products fall one or more octaves away fromthe desired signal, and can usually be satisfactorily filtered out witha low pass filter that blocks all frequencies above a pre-selectedcut-off frequency. However, if the receiver is a wide band or multioctave bandwidth receiver, these harmonics will fall within thebandwidth of the receiver and cannot be low pass filtered, without alsofiltering out some of the desired signals. In this case, other methodsknown to those skilled in the art, such as reducing the distortionproducts produced, must be used to eliminate this distortion.

Radio signals do not exist in isolation. The radio frequency spectrum ispopulated by many channels within a given band transmitting at variousfrequencies. When a radio circuit is presented with two or morefrequencies, these frequencies interact, or intermodulate, to createdistortion products that occur at known frequency locations.

FIG. 3 is an illustration of intermodulation distortion products.Whenever two or more frequencies are present they interact to produceadditional spurious signals that are undesired. FIG. 3 illustrates aspurious response produced from the interaction of two signals, f₁ 302and f₂ 304. This particular type of distortion is called intermodulationdistortion (IMD). These intermodulation distortion products 306 areassigned orders, as illustrated. In classifying the distortion the IMproducts are grouped into two families, even and odd order IM products.Odd order products are shown in FIG. 3.

In a narrow band systems the even order IM products can be easilyfiltered out, like harmonics, because they occur far from the twooriginal frequencies. The odd order IM products 306 fall close to thetwo original frequencies 302, 304. In a receiver these frequencies wouldbe two received signals or a received channel and a local oscillator.These products are difficult to remove. The third order products 306 arethe most problematic in receiver design because they are typically thestrongest, and fall close within a receiver's tuning band close to thedesired signal. IM distortion performance specifications are importantbecause they are a measure of the receiver's immunity to strong out ofband signal interference.

Third order products 308 occur at (f₁−Δf) and at (f₂+Δf), whereΔf=f₂−f₁. These unwanted signals may be generated in a transmitter andtransmitted along with desired signal or are created in a receiver.Circuitry in the receiver is required to block these signals. Theseunwanted spurious responses arise from nonlinearities in the circuitrythat makes up the receiver.

The circuits that make up the receiver though nonlinear are capable ofoperating linearly if the signals presented to the receiver circuits areconfined to signal levels within a range that does not call foroperation of the circuitry in the nonlinear region. This can be achievedby careful design of the receiver.

For example, if an amplifier is over driven by signals presented to itgreater than it was designed to amplify, the output signal will bedistorted. In an audio amplifier this distortion is heard on a speaker.In a radio receiver the distortion produced in nonlinear circuits,including amplifiers and mixers similarly causes degradation of thesignal output of the receiver. On a spectrum analyzer this distortioncan be seen; levels of the distortion increase to levels comparable tothe desired signal.

While unwanted distortion such as harmonic distortion, can be filteredout because the harmonics most often fall outside of the frequency bandreceived, other distortion such as inter-modulation distortion is moreproblematic. This distortion falls within a received signal band andcannot be easily filtered out without blocking other desired signals.Thus, frequency planning is often used to control the location ofdistortion signals that degrade selectivity.

Frequency planning is the selection of local oscillator signals thatcreate the intermediate frequency (IF) signals of the down conversionprocess. It is an analytical assessment of the frequencies being usedand the distortion products associated with these frequencies that havebeen selected. By evaluating the distortion and its strength, anengineer can select local oscillator and IF frequencies that will yieldthe best overall receiver performance, such as selectivity and imageresponse. In designing a radio receiver, the primary problemsencountered are designing for sufficient sensitivity, selectivity andimage response.

Selectivity is a measure of a radio receiver's ability to reject signalsoutside of the band being tuned by a radio receiver. A way to increaseselectivity is to provide a resonant circuit after an antenna and beforethe receiver's frequency conversion circuitry in a “front end.” Forexample, a parallel resonant circuit after an antenna and before a firstmixer that can be tuned to the band desired will produce a highimpedance to ground at the center of the band. The high impedance willallow the antenna signal to develop a voltage across this impedance.Signals out of band will not develop the high voltage and are thusattenuated.

The out of band signal rejection is determined by a quality factor or“Q” of components used in the resonant circuit. The higher the Q of acircuit in the preselector, the steeper the slope of the impedance curvethat is characteristic of the preselector will be. A steep curve willdevelop a higher voltage at resonance for signals in band compared tosignals out of band. For a resonant circuit with low Q a voltagedeveloped across the resonant circuit at a tuned frequency band will becloser in value to the voltage developed across the resonant circuit outof band. Thus, an out of band signals would be closer in amplitude to anin band signals than if a high Q circuit were constructed.

This type of resonant circuit used as a preselector will increasefrequency selectivity of a receiver that has been designed with thisstage at its input. If an active preselector circuit is used between anantenna and frequency conversion stages, the sensitivity of the receiverwill be increased as well as improving selectivity. If a signal is weakits level will be close to a background noise level that is present onan antenna in addition to a signal. If this signal cannot be separatedfrom the noise, the radio signal will not be able to be converted to asignal usable by the receiver. Within the receiver's signal processingchain, the signal's amplitude is decreased by losses at every stage ofthe processing. To make up for this loss the signal can be amplifiedinitially before it is processed. Thus, it can be seen why it isdesirable to provide a circuit in the receiver that provides frequencyselectivity and gain early in the signal processing chain.

Radio frequency tuners are increasingly being designed with majorportions of their circuitry implemented as an integrated circuit. In thestate of the art to minimize distortion products created in thereceiver, exotic materials such as gallium arsenide (GaAs) are used. Areceiver implemented on this type of material will typically have lowerdistortion and noise present than in a similarly constructed receiverconstructed on silicon. Silicon, is an attractive material due to itslow cost. In addition, a CMOS circuit implemented on silicon has theadditional benefit of having known processing characteristics that allowa high degree of repeatability from lot to lot of wafers. The state ofthe art has not achieved a completely integrated receiver in CMOScircuitry. A reason for this is the difficulty of eliminating receiverdistortion and noise.

The distortion products discussed above that are created in the receivercan, in the majority of cases, also be reduced by setting an appropriatedrive level in the receiver, and by allowing a sufficient spacingbetween carriers and channels. These receiver design parameters aredependent upon many other factors as well, such as noise present in thesystem, frequency, type of modulation, and signal strength among others.Noise is one of the most important of these other parameters thatdetermines the sensitivity of the receiver, or how well a weak signalmay be satisfactorily received.

Noise is present with the transmitted signal, and also generated withina receiver. If excessive noise is created in a receiver a weak signalmay be lost in a “noise floor”. This means that the strength of thereceived signal is comparable to the strength of the noise present, andthe receiver is incapable of satisfactorily separating a signal out ofthis background noise, or floor. To obtain satisfactory performance a“noise floor” is best reduced early in a receiver's chain of circuitcomponents.

Once a signal is acquired and presented to a receiver, in particularlyan integrated receiver with external pins, additional noise may beradiated onto those pins. Thus, additional added noise at the receiverpins can degrade the received signal.

In addition to the noise that is present on an antenna or a cable inputto a receiver, noise is generated inside the radio receiver. At a UHFfrequency range this internal noise predominates over the noise receivedwith the signal of interest. Thus, for the higher frequencies theweakest signal that can be detected is determined by the noise level inthe receiver. To increase the sensitivity of the receiver a“pre-amplifier” is often used after an antenna as a receiver front endto boost the signal level that goes into the receiver. This kind ofpre-amplification at the front end of the amplifier will add noise tothe receiver due to the noise that is generated inside of this amplifiercircuit. However, the noise contribution of this amplifier can beminimized by using an amplifier that is designed to produce minimalnoise when it amplifies a signal, such as an LNA. Noise does not simplyadd from stage to stage; the internal noise of the first amplifiersubstantially sets the noise floor for the entire receiver.

In calculating a gain in a series of cascaded amplifiers the overallgain is simply the sum of the gains of the individual amplifiers indecibels. For example, the total gain in a series of two amplifiers eachhaving a gain of 10 dB is 20 dB for a overall amplifier. Noise floor iscommonly indicated by the noise figure (NF) The larger the NF the higherthe noise floor of the circuit.

A cascaded noise figure is not as easily calculated as amplifier gain;its calculation is non-intuitive. In a series of cascaded amplifiers,gain does not depend upon the positioning of the amplifiers in thechain. However, in achieving a given noise figure for a receiver, theplacement of the amplifiers is critical with respect to establishing areceiver's noise floor. In calculating the noise figure for anelectronic system Friis' equation is used to calculate the noise figureof the entire system. Friis' equation is: $\begin{matrix}{{NF}_{total} = {{NF}_{1} + \frac{{NF}_{2} - 1}{G_{1}} + \frac{{NF}_{3} - 1}{G_{1}G_{2}} + \ldots + \frac{{NF}_{n} - 1}{G_{1}G_{2}\quad \ldots \quad G_{n}}}} & (1)\end{matrix}$

NF_(total)=system noise figure

NF₁=noise figure of stage-1

NF₂=noise figure of stage-2

NF_(n)=noise figure of stage-nth

G₁=gain of stage-1

G₂=gain of stage-2

G_(N)=gain of nth stage

What can be seen from this equation is that the noise figure of a firststage is the predominant contributor to a total noise figure. Forexample, the noise figure of a system is only increased a small amountwhen a second amplifier is used. Thus, it can be seen that the noisefigure of the first amplifier in a chain of amplifiers or systemcomponents is critical in maintaining a low noise floor for an entiresystem or receiver. A low NF amplifier typically requires a low noisematerial for transistors, such as gallium arsenide. Later amplifiersthat do not contribute significantly to the noise, are constructed of acheaper and noisier material such as silicon.

The initial low noise amplifiers are typically constructed fromexpensive materials such as gallium arsenide to achieve sufficientperformance. Gallium arsenide requires special processing, furtheradding to its expense. Additionally, GaAs circuits are not easilyintegrated with silicon circuits that make up the bulk of the receiversin use. It would be desirable to achieve identical performance with aless costly material, such as silicon. Silicon requires less costlyprocessing. Further it is advantageous if a standard process, such asCMOS, could be used to achieve the required low noise design. Given thetrend towards miniaturization and high volume production, it is highlydesirable to be able to produce an integrated receiver with a low noisefloor on silicon.

Within a receiver the layout and spacing of circuitry is critical toavoid the injection of noise generated in other portions of the circuitonto a received signal. If a tuner is placed on a semiconductorsubstrate noise generated in the substrate itself will interfere with,and degrade the received signal, this has been a problem preventingcomplete integration of a receiver on silicon.

Historically low noise substrates, fabricated from exotic and costlymaterials such as gallium arsenide have been used to reduce noisegenerated by the semiconductor substrate. However, it would beadvantageous to be able to fabricate a receiver on a single CMOSsubstrate. CMOS advantageously is a known process that may beimplemented economically for volume production. Currently a receiverfabricated completely in CMOS has not been available without utilizingexternal components in the received signal path. Each time the signal isrouted on or off of the integrated circuit additional opportunities forthe introduction of noise into a signal path are provided. Minimizingthis introduction of noise is an ongoing problem in receiver design.

After preselection and low noise amplification that is performed in afront end of a receiver, the signal next enters the receiver's frequencyconversion circuitry. This circuitry takes channels that have beenpassed through the front end and converts one of the selected channel'sfrequencies down to one or more known frequencies (f_(IF) or IFs). Thisfrequency conversion is accomplished through the use of a circuit calleda mixer that utilizes a local oscillator signal (f_(LO)), usuallygenerated in the receiver, to tune a received channel to an IF frequencywhile blocking the other channels. Spurious signals, previouslydescribed, are produced in this receiver circuitry, and an additionalproblem known as “image response” is encountered that must be consideredin the receiver's design.

It is well known to those skilled in the art that when two sinusoidalsignals of differing frequencies are multiplied together by theirapplication to a nonlinear device, such as a mixer, that signals of adiffering frequency are produced. A mixer has three ports: f_(RF)receives a low level radio frequency signal that contains the desiredmodulation, f_(LO) is a high level signal from a local oscillator, andf_(IF) is the resultant mixer product or intermediate frequencyproduced. These frequencies are related:

f _(IF) =mf _(RF) ±nf _(LO)  (2)

where

m=0, 1, 2, 3, . . . and

n=0, 1, 2, 3, . . .

In a typical first order circuit (m=n=1) four frequencies are produced:f_(RF), f_(LO), f_(IFLO)=f_(RF)−f_(LO) and f_(IFHI)=f_(RF)+f_(LO). Af_(IFLO) and f_(IFHI) being termed intermediate frequencies. Inreceivers the common practice is to select either the sum or differenceIF frequency by filtering out the undesired one. Since both signalscontain the same information, only one is needed in the subsequentcircuitry.

One or more mixers are advantageously used in radio receivers to converta high frequency radio signal which is received into a lower frequencysignal that can be easily processed by subsequent circuitry. Mixers arealso used to tune multiple channels, so that different tuned circuitsare not required for each channel. By changing a local oscillatorfrequency, differing radio frequencies received can be tuned to producea constant intermediate frequency value regardless of the frequency ofthe received channel. This means that circuit components used to processthe intermediate frequency may be fixed in value, with no tuning ofcapacitors or coils required. Thus, circuits in an IF strip are allfixed-tuned at an IF frequency. A receiver constructed in this manner,using one or more frequency conversions, is called a superheterodyneradio receiver.

A disadvantage of a superheterodyne radio receiver is that any of theone or more local oscillators within the receiver also acts as aminiature transmitter. A receiver “front end” alleviates this problem byisolating an antenna from the remaining receiver circuitry.

By positioning a radio frequency amplifier between the antenna and thefrequency converting stages of a receiver, additional isolation betweenthe receiver circuitry and the antenna is achieved. The presence of anamplifier stage provides attenuation for any of the one or more localoscillator signals from the frequency conversion stages that areradiated back towards the antenna or a cable distribution network. Thisincreased isolation has the benefit of preventing radiation of a localoscillator signal out the antenna which could cause radio frequencyinterference from a local oscillator. If radiated these and othersignals present could create interference in another receiver present atanother location.

FIG. 4 is an illustration that shows an image frequency's 402 relationto other signals present 404, 406, 408 at a mixer. Image frequencysuppression is an important parameter in a receivers design. In a radioreceiver two frequencies input to a radio receiver 404, 406 will yield asignal at the IF frequency 408. A receiver will simultaneously detectsignals at the desired frequency 404 and also any signals present at anundesired frequency known as the image frequency 402. If there is asignal present at the image frequency, it will translate down to the IFfrequency 408 and cause interference with the reception of the desiredchannel. Both of these signals will be converted to the IF frequencyunless the receiver is designed to prevent this. The image frequency 402is given by:

f _(I) =f _(RF)+2f _(IF)  (3)

where f_(I) is the image frequency. This is illustrated in FIG. 4. Afrequency that is spaced the IF frequency 410 below the local oscillatorfrequency (f_(RF)) 404, and a frequency that is spaced the intermediatefrequency 412 above the local oscillator signal (f_(I)) 402, will bothbe converted down to the intermediate frequency (f_(IF)) 408. The usualcase is that a frequency that occurs lower than the local oscillatorsignal is the desired signal. The signal occurring at the localoscillator frequency plus the intermediate frequency 402 is an unwantedsignal or noise at that frequency that is converted to the IF frequencycausing interference with the desired signal.

In FIG. 4 the exemplary 560 KHz signal 404 is a radio station that thetuner is tuned to receive. The exemplary 1470 KHz signal 402 is anotherradio station transmitting at that particular frequency. If a designerof the receiver had picked an exemplary local oscillator signal of 1015KHz 406 then both of these radio stations would be simultaneouslyconverted to an exemplary IF frequency of 455 KHz 408. The personlistening to the radio would simultaneously hear both radio programscoming out of his speaker. This illustrates the need for the carefulselection of local oscillator frequencies when designing a radioreceiver. The selection of local oscillator frequencies is a part offrequency planning and used by those skilled in the art to design areceiver that will provide frequency conversions needed with minimaldistortion.

FIG. 5 illustrates a dual (or double) conversion receiver 502. Such amultiple conversion receiver allows selectivity, distortion andstability to be controlled through a judicious frequency planning. Inthe double conversion receiver 502 a received signal 504 is first mixed506 to a first intermediate frequency, and then mixed 508 down to asecond intermediate frequency. In this type of receiver the first IFfrequency is made to be high so that a good image rejection is achieved.The second IF is made low so that good adjacent channel selectivity isachieved.

If the first IF frequency is low an image frequency falls higher infrequency, or closer to the center of a pass band of an RF selectivitycurve of a receiver “front end,” 510 and undergoes little attenuation.If the IF frequency is high the image frequency falls far down on theskirt of the RF selectivity curve for the receiver “front end” receivinga required attenuation. Thus, the selectivity of the receiver acts toattenuate the image frequency when a high IF frequency is used. As anadded benefit a high image frequency provides less of a chance forinterference from a high powered station. This is because at higherfrequencies transmitted power is often lower due to the difficulties ingenerating RF power as frequency increases.

A low second IF frequency produces a good adjacent channel selectivity.Frequency spacing between adjacent channels is fixed. To preventinterference from adjacent channels the receiver must possess a goodselectivity. Selectivity can be achieved through a RF tuned circuit, andmore importantly by the superior selectivity provided by a frequencyconversion process. The selectivity improvement given by using a low IFis shown by considering a percent separation of a desired and anundesired signal relative to total signal bandwidth. If a separationbetween the desired and undesired signals is constant a second IF signalfalling at the lower frequency will give a larger percent separationbetween the signals. As a result it is easier to distinguish between IFsignals that are separated by a larger percentage of bandwidth. Thus,the judicious selection of two intermediate frequencies in a doubleconversion receiver is often used to achieve a given design goal, suchas image frequency rejection and selectivity.

Additionally, the use of a second IF frequency allows gain in thereceiver to be distributed evenly. Distributing gain helps preventinstability in the receiver. Instability usually is seen as anoscillating output signal 512. Distributing the gain among several IFamplifiers 514, 516, 518 reduces the chance of this undesirable effect.Often to further distribute the gain required in a system design a thirdfrequency conversion, and a third IF frequency, will be utilized.

After a receiver front end that possibly contains a low noise amplifier,additional amplifiers are often seen in the various IF strips. Anamplifier in an IF strip does not require frequency tuning and providessignal gain to make up for signal losses, encountered in processing areceived signal. Such losses can include conversion loss in mixers andthe insertion loss encountered by placing a circuit element, such as afilter or an isolator in the IF strip.

In receivers filters are used liberally to limit unwanted frequenciesthat have been escaped previous elimination in a “front end,” or toeliminate unwanted frequencies that have been created immediatelypreceding a filter. In addition to attenuating unwanted frequencies, adesired signal will also undergo some attenuation. This attenuationresults from an insertion loss of a filter, or some other component, andif uncompensated, will degrade a signal. This is especially true when aseries of filters are cascaded, since the effect is additive.

Often a series of multiple filters are cascaded in a given IF strip.These filters typically have an identical response characteristic. Thecascaded filters are used to increase the selectivity of the receiver.While it is true that the insertion loss in the pass band is the sum ofindividual filter insertion losses, as measured in decibels, a rejectionimprovement obtained outside of the pass band is the sum of therejections at the given frequency. Thus, three cascaded filters, eachhaving an insertion loss of 0.01 dB at a center frequency, would have atotal insertion loss of 0.03 dB. If the rejection in the stop band, agiven frequency away from the center frequency of the filter, were 20dB, then a total rejection for 3 cascaded filters would be 60 dB, agreat improvement in filter selectivity.

In choosing intermediate frequencies for IF strips in the receiver, noconcrete design guidelines exist. Also because of a wide variance indesign goals that are encountered in receiver design, concretemethodologies do not exist. Each receiver must be uniquely engineered tosatisfy a series of system design goals taking into consideration designtradeoffs that must be made. In the current state of the art, designtradeoffs, and design methodologies used have been directed tointegrating all parts of the receiver except for frequencies selectivecomponents. The conventional wisdom in receiver design is that filtersare not easily integrated onto a silicon substrate and that filtering isbest done off of a chip.

Some general design guidelines exist to aid an RF engineer in designinga receiver. One such rule is that designing for receiver selectivity ismore important than designing for receiver sensitivity. Thus, when facedwith conflicting design choices, the more desirable choice is to providea design that will separate adjacent channels that interfere with eachother rather than to design a receiver capable of picking up the weakestchannels. Another rule of thumb in choosing intermediate frequencies isto choose the first intermediate frequency at twice the highest inputfrequency anticipated. This is to reduce the possibility of spurioussecond order intermodulation distortion. Depending upon a systemperformance desired, this rule can even be more restrictive, requiringan IF at greater than three times the highest input frequency. Thus, itmay be seen that a wide variety of performance requirements exist in areceiver circuit, and that the range of choices for a given criteria maybe utilized by those skilled in the art to produce a unique design thatmeets the challenges posed by an increasing trend towards integration.

When more than one IF is present in a receiver there is an imagefrequency associated with each IF that must be considered in the design.A good receiver provides an image rejection greater than 70 dB.

One of the first considerations in frequency planning a superheterodynereceiver is the selection of IF conversions. A frequency range of thelocal oscillator needs to be determined to establish the locations ofspurious responses of various orders. Two choices are possible for eachof two possible LO frequency and the selection is not subject to an easygeneralization. The two available frequencies are the absolute value ofthe quantity |f_(RF)±f_(IF)|=f_(LO). Selection depends on RF bandschosen to be received and frequencies present in these bands, theavailability of fixed bandwidth filters at a desired IF and constraintsimposed upon an engineer by the limitations of a material that will beused to fabricate a receiver.

Receiver planning is a process that is centered upon frequency planningand receiver level diagrams. After initial frequency selections for afrequency plan are made, a receiver level plan is used to calculatenoise figures, intercept points (IP) and levels of spurious responses.Each is evaluated in light of design requirements. After each set ofselections performance is evaluated and a next set of parameterselections is made until an appropriate compromise in receiverperformance is achieved.

Once frequency planning and a level diagram yield a satisfactory designsolution these tools are used to guide a detailed receiver design. Onceparameters of a section of a receiver are defined, an engineer can usevarious circuit implementations to achieve a stated design goal. Forexample a frequency plan and level diagram may require a band passfilter with certain characteristics such as bandwidth, center frequencyand insertion loss. The engineer would then either pick a single filterthat meets all of these requirements or cascade one or more filters suchthat a composite response will yield the required design value.

Needless to say experience and knowledge of available technology plays alarge part in achieving a successful receiver design blueprint. Anengineer must have a rough idea of component availability and designmethodologies that will yield a certain performance. If the engineerspecifies a portion of the receiver that has performance characteristicsthat are not achievable with available components or design methods,then an impractical and unproduceable design has been proposed requiringreplanning the architecture of the receiver.

A design process and a result achieved is very dependent upon technologyavailable, materials and methodologies known at the time. Newimprovements in design techniques, computer simulation, processing and apush for increased miniaturization continually fuel achievement of newand innovative receiver designs to solve technological problems.

Once frequency conversions have been chosen and a receiver designed,with the distortion products created in the receiver found acceptable,the next step in receiver design is to design circuitry that willgenerate one or more local oscillator signals. These signals could beprovided by a source that is external to a chip. However, this would notbe practical in seeking to miniaturize an overall receiver design. Abetter approach is to generate the local oscillator frequencies near thereceiver. In reducing an entire receiver onto a single chip, problems inmaintaining signal purity, and stability are encountered.

An innovation that has allowed increased miniaturization in receiverdesign is the development of frequency synthesis. Local oscillatorsignals are required in receivers utilizing frequency conversion. Thesesignals must be tunable and stable. A stable frequency is easilyproduced by a quartz crystal at a single frequency. A tunable frequencycan be produced by an LC type oscillator. However, this LC oscillatordoes not have sufficient stability. Additionally using a large number ofcrystals to generate a range of local oscillator signals, or inductorsrequired in an LC oscillator do not allow an easily miniaturized design.Frequency synthesis is space efficient.

Variable frequency local oscillator signals used in a receiver must begenerated by appropriate circuits. These frequency synthesis techniquesderive variable LO signals from a common stable reference oscillator. Acrystal oscillator has a stable frequency suitable for use in asynthesizer.

Oscillators may provide a fixed or a variable output frequency. Thisfixed or variable frequency may be used for frequency conversion in areceiver as a local oscillator that is used to mix a received radiofrequency (RF) input down to an intermediate frequency or a base bandsignal that is more easily processed in the following circuitry. Anotherway that a received signal can be converted down to a base band orintermediate frequency signal is by using frequency synthesizer outputsas local oscillator signals to mix the signal down. Synthesizers provideaccurate, stable and digitally programmable frequency outputs, withoutthe use of multiple oscillators to tune across a band. Accuracy ismaintained by using feed back.

Three general techniques are used for frequencies synthesis. Directsynthesizers use frequency multipliers, dividers and mixers. Indirectsynthesizers use phase-locked loops. Direct digital synthesizers usedigital logic combined with a digital to analog converter to provide ananalog output. Some designs combine the three techniques.

A direct synthesizer will use a frequency reference such as a crystaloscillator as disclosed in FIG. 5 to generate a reference frequency. Toachieve a desired output frequency, the reference frequency ismultiplied through a series of multipliers. Dividers may be usedsimilarly to reduce the frequency output to the desired lesser value.Additionally, two signals generated from the chain of multipliers anddividers can be fed into a mixer to generate a third frequency. The mixand divide direct synthesis approach permits the use of many identicalmodules that produce fine resolution with low spurious output.

Indirect synthesis can take several forms. It can use divide by N toproduce one or more of the digits, and mix and divide with loopsimbedded among circuits. In each form of frequency synthesizer, theloops contained in it are governed by a derivative of a referencefrequency. Indirect synthesis can be used to generate a frequency of(N/M) f_(in). Circuits of this type are often used as local oscillatorsfor digitally tuned radio and television receivers.

Indirect synthesizers make use of a number of phase locked loops (PLLs)in order to create a variety of frequency outputs. Each loop present inthe system makes use of a common frequency reference provided by asingle oscillator. Frequency synthesizers provide the advantage of beingdigitally programmable to a desired frequency as well as providing anextremely stable frequency.

Frequency stability in a synthesizer is achieved with phase lockedloops. A phase locked loop is programmed to generate a desiredfrequency. Once it approximates the frequency, the frequency is divideddown to the value of a reference frequency, provided by an externaloscillator, and compared to that reference frequency. When thedifference reaches zero the phase locked loop stops tuning and locks tothe frequency that it has just produced. The frequency reference used totune the phase locked loop is typically provided by a single frequencyoscillator circuit.

Frequency synthesizers in a radio frequency receiver often incorporatetwo phase locked loops. One PLL is used to provide coarse tuning withinthe frequency band of interest while the second PLL provides fine tuningsteps.

In using this scheme, a coarse tuning must be such that a desiredchannel will initially fall within the selectivity of the receiver toproduce a signal output. It would be an advantage in receiver design iftuning speed could be increased so that initially several channels wouldfall within the selectivity of the receiver. Tuning in this manner wouldallow an output to be created with an extremely coarse tuning range thatcould be dynamically adjusted. Currently this type of tuning is not seenin the state of the art.

Typically PLLs use a common reference frequency oscillator. Localoscillator signals produced by a frequency synthesizer's phase lockedloops inject noise produced in the reference frequency oscillator andthe PLLs into a the signal path by way of a PLL output.

A range of output frequencies from a synthesizer can span many decades,depending on the design. A “resolution” of the synthesizer is thesmallest step in frequency that can be made. Resolution is usually apower of 10. A “lock up time” of the synthesizer is the time it takes anew frequency to be produced once a command has been made to changefrequencies.

The more accurate the frequency required the longer the lock up time.The reduction of the lock up time is a desirable goal in synthesizerdesign. A modern trend is to use frequency synthesis in wide bandtuners. To tune across a wide band width quickly the lock up time mustbe minimized. Current state of the art tuning times for jumps infrequencies can be as short as several microseconds. This is difficultto do when the required increment in frequency adjustment is small. Inthe state of the art indirect synthesis is capable of producing multidigit resolution. However, indirect synthesis is not capable ofproviding micro second switching speeds. For faster switching speedsdirect analog and direct digital technologies are used. Therefore, it isdesirable to construct an indirect frequency synthesizer that provideshigh resolution and improved switching speed.

The present embodiments of the invention allow all channel selectivityand image rejection to be implemented on an integrated circuit.Integration is a achievable by utilizing differential signaltransmission, a low phase noise oscillator, integrated low Q filters,filter tuning, frequency planning, local oscillator generation and PLLtuning to achieve a previously unrealized level of receiver integration.

The embodiments of the invention advantageously allow a LC filters to beintegrated on a receiver chip, resulting in an integrated circuit thatcontains substantially the entire receiver. By advantageously selectinga frequency plan, and utilizing the properties of complex mixers, anarchitecture is achieved that allows LC filters to be integrated on areceiver chip so that acceptable performance is produced when convertinga received signal to one having a lower frequency that is easilyprocessed.

The embodiments utilize particular aspects of an arbitrarily definedinput spectrum to first shift the received frequencies to a higherfrequency in order that interference may be more easily eliminated byfiltering and then shifting the spectrum to a nominal IF for processing.This first shifting process advantageously shifts interfering imagesignals away from a center frequency of a first LC filter bank so thatthe LC filter bank is more effective in reducing the interfering signalstrength. To further reduce the interfering signal strength, multiple LCfilters that are tuned to the same frequency are cascaded, furtherreducing the interfering signal strength.

To reduce degradation of the desired signal the exemplary embodiments ofthe invention utilize a complex mixing stage following an LC filter bankto reduce the image frequency interference by an additional amount thatmight be necessary to meet a particular image rejection target (i.e., anabout 60 dB to 65 dB rejection target). A complex mixer creates a signalas a result of its normal operation that cancels an image frequencyinterference by the remaining amount needed to achieve satisfactoryperformance with LC filters.

The ultimate goal of a receiver is to reduce the frequency of anincoming signal to a frequency that is lower than received, so thatprocessing of the desired signal can be easily achieved. The receiverarchitecture utilizes two frequency down conversions to achieve thisgoal. Each frequency conversion is susceptible to interference thatrequires filtering. Frequency planning as described above used inconjunction with LC filters and complex mixers, provides the requiredimage distortion rejection that allows LC filters to be usedadvantageously in an integrated receiver.

Radio receivers require one or more local oscillator (LO) signals inorder to accomplish frequency conversion to an intermediate (IF)frequency. In a typical receiver these local oscillator signals must bestable and free from noise. When a receiver is fabricated as anintegrated circuit, the chances of injecting noise via the LO signalsincreases. Local oscillator signals for a receiver are typicallygenerated in close proximity to the frequency conversion circuitry. Theclose proximity of this frequency generation circuitry to the signalpath creates an increased likelihood of noise being radiated orconducted to cause interference with the received signal.

In order to achieve improved noise immunity the exemplary embodiments ofthe invention may utilize circuitry to generate the local oscillatorsignals that possess superior noise performance. The local oscillatorsignals may also be advantageously transmitted differentially to themixers present on the integrated circuit. It should be noted that inalternate embodiments of the invention that a single ended output can beproduced from the differential signal by various techniques known in theart. This technique is used advantageously whenever external connectionsto the receiver are required that are single ended.

Oscillator

An exemplary embodiment of the present invention utilizes a differentialoscillator having low phase noise or jitter and high isolation, as afrequency reference that substantially increases the performance of atuner architecture integrated onto a single silicon substrate.

In accordance with the present invention, a crystal oscillator circuitis provided and constructed so as to define a periodic, sinusoidal,balanced differential signal across two symmetrical terminals of acrystal resonator which are coupled in a parallel configuration acrosssymmetrical, differential terminals of a differential oscillatorcircuit.

The differential oscillator circuit is configured such that it isconstructed of simple active and passive components which are easilyimplemented in modern integrated circuit technology, thus allowing thedifferential oscillator circuit to be accommodated on a monolithicintegrated circuit chip for which the crystal oscillator (as a whole) isproviding a suitable, stable periodic timing reference signal.Similarly, and in contrast to prior art implementations, only theresonating crystal (crystal resonator or quartz crystal resonator) isprovided as an off-chip component. This particular configuration allowsfor considerable savings in component parts costs by partitioning moreand more functionality into the integrated circuit chip.

Remote (off chip) mounting of the crystal resonator requires thatelectrical contact between the crystal resonator and the associatedoscillator circuit, be made with interconnecting leads of finite length.In integrated circuit technology, these interconnecting leads aretypically implemented as circuit pads and conductive wires formed on aPC board substrate to which package leads are bonded (soldered) in orderto effect electrical connection between the crystal resonator and anassociated oscillator circuit. External electrical connections of thistype are well known as being susceptible to noise and other forms ofinterference that might be radiated onto the interconnecting leads and,thence, into the oscillator circuit, degrading its overall noiseperformance.

A sinusoidal signal source, having a differential output configuration,defines a pair of periodic sinusoidal signals, with the signal at oneoutput terminal defined as being 180° out of phase with a similarperiodic, sinusoidal signal appearing at the other output terminal.Classical differential signals are termed “balanced” in that bothsignals exhibit equal peek-to-peek amplitudes although they exhibit a180° phase relationship. As illustrated in the simplified timing diagramof FIG. 6, differential signals have a particular advantage in thatcommon-mode interference, that is injected on either terminal, iscanceled when the signal is converted to single-ended. Such common modeinterference is typically of equal amplitude on each pin and is causedby radiation into the circuit from external sources or is oftengenerated in the circuit itself. In FIG. 6, a positive sinusoidalsignal, denoted signal-P oscillates about a zero reference, but isshifted by a common-mode interference component, denoted I_(CM).Likewise, a negative sinusoidal signal, denoted at signal-n, alsooscillates about a zero reference, exhibiting a 180° phase relationshipwith signal-p, and is also offset by a common mode interferencecomponent denoted I_(CM).

A superposition of the positive and negative periodic signals isillustrated in the timing diagram denoted “composite”, which clearlyillustrates that the peek-to-peek difference between the positive andnegative signals remains the same, even in the presence of a common modeinterference component I_(CM).

Turning now to FIG. 7, there is depicted a semi-schematic block diagramof a periodic signal generation circuit including a differential crystaloscillator driving a differential linear buffer amplifier.Advantageously, the present invention contemplates differential signaltransmission throughout its architecture to maintain the purity of thederived periodic signal and to minimize any common mode interferencecomponents injected into the system. In particular, the presentinvention incorporates differential signal transmission in theconstruction of a differential crystal oscillator circuit, including acrystal resonator and its associated oscillator driver circuit.Differential signal transmission is maintained through at least a firstlinear buffer stage which functions to isolate the differentialoscillator circuit switch transients and other forms of noise that mightbe generated by follow-on digital integrated circuit components.

In FIG. 7, a differential crystal oscillator circuit is configured tofunction as a source of stable, synchronous and periodic signals.According to the illustrated embodiment, a differential crystaloscillator 710 suitably incorporates a resonating crystal 712 and a pairof symmetrical load capacitors 714 and 716, each load capacitorrespectively coupled between ground potential and one of the twosymmetrical output terminals of the resonating crystal 712.

Resonating crystal 712 is coupled between differential terminals of adifferential oscillator driver circuit 718, in turn connected todifferential inputs of a differential linear buffer integrated circuit720. The symmetrical terminals of the resonating crystal 712 are coupledacross differential terminals of the resonator and linear buffer, with afirst terminal of the crystal being shunted to ground by the first shuntcapacitor 14. The second terminal of the crystal is shunted to ground bythe second shunt capacitor 716.

The oscillator driver circuit portion of the differential crystaloscillator 710 functions, in cooperation with the crystal resonator 712,to define a pure sinusoidal and differential signal across the crystal'ssymmetrical terminals. As will be developed in greater detail below,this pure sinusoidal and differential signal is then used by the linearbuffer 720 to develop an amplified representation of periodic signalssynchronized to the crystal resonant frequency. These amplified signalsare also contemplated as differential inform and are eminently suitablefor driving digital wave shaping circuitry to define various digitalpulse trains useable by various forms of digital timing circuitry, suchas phase-lock-loops (PLLs), frequency tunable digital filters, directdigital frequency synthesizers (DDFS), and the like. In other words, thesystem depicted in FIG. 7 might be aptly described as a periodicfunction generator circuit, with the crystal oscillator portion 710providing the periodicity, and with the buffer portion 720 providing thefunctionality.

Before entering into a detailed discussion of the construction andoperation of the differential oscillator driver circuit and differentiallinear buffer amplifier, it will be useful to describe characteristicsof a resonating crystal, such as might be contemplated for use in thecontext of the present invention.

FIG. 8 depicts the conventional representation of a resonating crystal712 having mirror-image and symmetrical terminals 822 and 824, uponwhich differential periodic signals may be developed at the crystal'sresonant frequency. Resonating crystals (also termed crystal resonators)may be formed from a variety of resonating materials, but most commonlyare formed from a piece of quartz, precisely cut along certain of itscrystalline plane surfaces, and so sized and shaped as to define aparticular resonant frequency from the finished piece. Resonatingcrystals so formed are commonly termed “quartz crystal resonators”.

A typical representational model of the equivalent circuit of a quartzcrystal resonator 712 is illustrated in simplified, semi-schematic formin FIG. 9. A quartz crystal resonator can be modeled as a two terminalresonator, with an LCR circuit, incorporating a capacitor C_(m) inseries with an inductor L_(m) and a resistor R_(m), coupled in parallelfashion with a capacitor C_(o) across the two terminals. It will beunderstood that the particular component values of the capacitor,inductor and resistor, forming the LCR filter portion of the circuit,define the resonant characteristics of the crystal. These design valuesmay be easily adjusted by one having skill in the art in order toimplement a resonating crystal operating at any reasonably desiredfrequency.

For example, a particular exemplary embodiment of a crystal resonatormight be desired to have a resonant frequency in the range of about 10megahertz (MHz). In such a case, the equivalent circuit of such acrystal might have a typical value of about 20 femto Farads (fF) for thecapacitor C_(m). The inductor L_(m) might exhibit a typical value ofabout 13 milli Henreys (mH), while the resistor might have a typicalvalue of about 50 ohms. When used in a practical oscillator design,oscillation will be achieved for values of the capacitor C₀ that areless than a design worst case value. In the exemplary embodiment, worstcase values of 7 pico Farads (pF) might be chosen in order to ensure adesign that oscillates at the desired resonant frequency over a widerange of crystal equivalent circuit values. In a practical application,the typical range of capacitance values for C₀ might be from about 3 toabout 4 pF.

FIGS. 10 and 11 are graphical representations depicting response plotsof impedance and phase with respect to frequency, respectively, of acrystal resonator circuit constructed in accordance with the equivalentcircuit model of FIG. 9 and using the values given above for thecomponent C_(m), L_(m), R_(m), and C₀ parts. FIG. 10 is a plot of thereal portion of impedance, in ohms, as a function of the resonator'sfrequency and mega Hertz. FIG. 11 is a representational plot of theimaginary impedance component (expressed as phase), again expressed as afunction of frequency in mega Hertz. From the representational plots, itcan be understood that an exemplary crystal resonator constructed inaccordance with the above values exhibits a resonant frequency in therange of about 10 MHz. Further, simulation results on such a crystalresonator exhibit a steep rise in the real impedance versus frequencyplot of FIG. 10 in the resonance region about 10 MHz. A steep rise inreal impedance in the resonance region is indicative of a high qualityfactor, Q, typically exhibited by quartz crystal resonators.

An example of a quartz crystal resonator having the aforementionedcharacteristics and exhibiting a resonance fundamental at about 10 MHzis a Fox HC49U, quartz crystal resonator, manufactured and sold by FoxElectronics of Ft. Myers, Fla. It should be noted, however, that thespecific values of a quartz crystal resonator, including its resonantfrequency, are not particularly important to practice of principles ofthe invention. Any type of crystal resonator may be used as theresonator component 712 of FIG. 7, so long as it is constructed withgenerally symmetrical terminals which can be driven, in a manner to bedescribed in greater detail below, by an oscillator driver circuit 718of FIG. 7 so as to develop a differential, sinusoidal signal withrespect to the two terminals. Further, the resonator need not oscillateat a frequency of 10 MHz. The choice of resonant frequency is solely afunction of a circuit designer's preference and necessarily depends onthe frequency plan of an integrated circuit in which the system of theinvention is used to provide periodic timing signals.

Turning now to FIG. 12, there is depicted a simplified schematic diagramof a differential oscillator driver circuit, indicated generally at 718,suitable for differential coupling to a crystal resonator in order todevelop balanced, differential sinusoidal signals for use by downstreamcomponents.

In the exemplary embodiment of FIG. 12, the differential oscillatordriver circuit 718 is constructed using common integrated circuitcomponents and is symmetrical about a central axis. The oscillatordriver 718 is constructed with a pair of P-channel transistors 1226 and1228 having their source terminals coupled in common and to a currentsource 1230 connected, in turn, between the common source terminals anda positive supply potential V_(DD). The gate terminals of each of theP-channel transistors 1226 and 1228 are coupled to the drain nodes ofthe opposite transistor, i.e., the gate terminal of P-channel transistor1228 is coupled to the drain node of P-channel transistor 1226, and viceversa.

Output terminals are defined at each of the transistor's drain nodes,with the drain node of P-channel transistor 1226 defining the “negative”terminal (Von) and the drain terminal of P-channel transistor 1228defining the “positive” output (Vop). Thus, it will be understood thatthe circuit is able to operate differentially by cross coupling thetransistors 1226 and 1228 in order to provide feedback.

Because transistors exhibit some measure of gain at all frequencies,particularly DC, conventional cross coupled transistors are oftenimplemented as latches in digital circuit applications where large DCcomponents are present. In the differential oscillator driver circuit718 of the invention, latching is prevented by removing the DC gaincomponent, while retaining the system's high frequency gain,particularly gain in the desirable 10 MHz region.

In order to substantially eliminate the gain component at lowfrequencies, a high pass filter is interposed between the gate andoutput terminals of each symmetrical half of the circuit. In particular,a high pass filter 1232 is coupled between the “negative” outputterminal and the gate terminal of P-channel transistor 1228. Likewise,the high pass filter 1234 is coupled between the “positive” outputterminal and the gate terminal of P-channel transistor 1226. Further,each of the high pass filters 1232 and 1234 are coupled between avirtual ground, identified as Vmid and indicated in phantom in theexemplary embodiment of FIG. 12, and the corresponding gate terminal ofthe respective one of the differential pair P-channel transistors 1226and 1228. Each of the high pass filters 1232 and 1234 are implemented asRC filters, each including a resistor and capacitor in a series-parallelconfiguration. Each capacitor is series-connected between an outputterminal and the gate terminal of a corresponding differential pairtransistor, while each resistor is coupled between a gate terminal andthe virtual ground. Thus, the first high pass filter 1232 includes acapacitor 1236 coupled between the “negative” terminal and the gateterminal of P-channel transistor 1228. A resistor 1238 is coupledbetween the gate of P-channel transistor 1228 and virtual ground.Similarly, the second high pass filter 1234 includes a capacitor 1240coupled between the “positive” terminal and the gate terminal ofP-channel transistor 1226. A resistor 1242 is coupled between the gateof P-channel transistor 1226 and the virtual ground.

In operation, high pass filter 1232 filters the input from Von prior toapplying that signal to the gate of its respective differential pairtransistor 1228. In like manner, high pass filter 1234 filters the inputfrom Vop prior to applying that signal to the gate of its respectivedifferential pair transistor 1226. Each of the high pass filters aresymmetrically designed and have component values chosen to give cutofffrequencies in the range of about 5 MHz. For example, filter capacitors1236 and 1240 might have values of about 1.5 pF, and filter resistors1238 and 1242 might have values in the range of about 718 Kohms. Whichwould give a filter yielding the desired 5 MHz cutoff. It will be thusunderstood that the differential oscillator driver circuit 18 will havenegligible gain at DC, while exhibiting its design gain values in thedesired region of about 10 MHz.

It should be understood that the component values for high pass filters1232 and 1234 were chosen to give a particular cut off frequency ofabout 5 MHz, allowing the oscillator driver circuit to exhibit fulldesign gain at a resonate frequency of about 10 MHz. If the resonantfrequency of the crystal oscillator circuit were required to have adifferent value, the components of the high pass filters 1232 and 1234would necessarily take on different values to accommodate the differentoperational characteristics of the circuit. Accordingly, the actualcomponent values, as well as the cutoff frequency value of the exemplaryembodiment, should not be taken as limiting the differential oscillatordriver circuit according to the invention in any way. The values andcharacteristics of the differential oscillator driver circuit 18 of FIG.12 are exemplary and have been chosen to illustrate only one particularapplication.

Because the common mode output signal of a differential amplifier isoften undefined, the differential oscillator driver circuit 718 of FIG.12 is provided with a common mode control circuit which functions tomaintain any common mode output signal at reasonable levels. Inparticular, a differential pair of N-channel transistors 1244 and 1246is provided with each having its drain terminal coupled to a respectiveone of the Von and Vop output terminals. The differential N-channeltransistors 1244 and 1246 further have their source terminals tiedtogether in common and to a negative supply potential V_(ss). Their gateterminals are tied together in common and are further coupled, infeedback fashion, to each transistor's drain node through a respectivebias resistor 1248 and 1250. The bias resistors 1248 and 1250 each havea value, in the exemplary embodiment, of about 100 Kohms, with the gateterminals of the N-channel differential pair 1244 and 1246 coupled to acenter tab between the resistors. This center tab defines the virtualground Vmid which corresponds to a signal midpoint about which thesinusoidal signals Von and Vop oscillate. Any common mode componentpresent at the outputs will cause a voltage excursion to appear at thegates of the N-channel differential pair 1244 and 1246. In other words,virtual ground Vmid can be thought of as an operational threshold forthe current mode control differential pair 1244 and 1246. Common modeexcursions above or below Vmid will cause a common mode controldifferential pair to adjust the circuit's operational characteristics soas to maintain Vmid at a virtual ground level, thus minimizing anycommon mode component.

In operation, noise in such a linear differential oscillator drivercircuit is filtered mainly by the crystal resonator, but also by theoperational characteristics of the driver circuit. For example, noise at10 MHz is amplified by the positive feedback characteristics of thecircuit and will continue to grow unless it is limited. In the exemplaryembodiment of FIG. 12, signals in the 10 MHz region will continue togrow in amplitude until limited by a non-linear self-limiting gaincompression mechanism.

As the amplitude of the amplified signal becomes large, the effectivetransconductance g_(m) of the P-channel differential pair transistors1226 and 1228 fall off, thus limiting the gain of the differentialamplifier. Amplifier gain falloff with increasing gate voltageexcursions is a well understood principle, and need not be described inany further detail herein. However, it should be mentioned that as thegain of the oscillator driver circuit trends to 1 the crystal resonatorbegins to self-limit, thus defining a constant output amplitudesinusoidal signal. Constancy of the amplitude excursions are reflectedto the control (gate) terminals of the P-channel differential pair 1226and 1228 where the feedback mechanism ensures stability about unitygain.

It should be understood therefore that the differential oscillatordriver circuit 718 in combination with a crystal resonator (712 of FIG.7) function to define periodic, sinusoidal and differential signalsacross the terminals of the crystal resonator. The signals aredifferential in that they maintain a 180° phase relationship. Signalquality is promoted because the exemplary differential oscillator drivercircuit is designed to be highly linear with a relatively low gain, thusreducing phase noise (phase jitter) to a significantly better degreethan has been achieved in the prior art. Signal quality and symmetry isfurther enhanced by the symmetrical nature of the two halves of theoscillator driver circuit. Specifically, the oscillator driver circuitis symmetrical about a central axis and, when implemented in integratedcircuit technology, that symmetry is maintained during design andlayout. Thus, conductive signal paths and the spatial orientation of thedriver's active and passive components are identical with respect to the“negative” and “positive” outputs, thereby enhancing signal symmetry andfurther minimizing phase jitter.

In accordance with the invention, differential crystal oscillatorcircuit is able to provide a periodic clock signal (approximately 10MHz) that exhibits stable and robust timing characteristics with verylow jitter. As depicted in the simplified semi-schematic block diagramof FIG. 13, a particular exemplary embodiment of a periodic signalgenerator circuit incorporates a differential crystal oscillator circuitaccording to the present invention, including a crystal resonator 12 anddifferential oscillator driver circuit 718. A resonant crystal circuit12 includes first and second timing capacitors (714 and 716 of FIG. 7)which are not shown merely for convenience in ease of explanation. Theresonant crystal circuit 712 is coupled, in parallel fashion, across theoutput terminals of the oscillator driver circuit 718 which incorporatesthe active device circuitry for pumping energy into the circuit in orderto sustain oscillation. This parallel combination is coupled,differentially, into a linear buffer amplifier 720, which functions toprovide a linear gain factor K to the differential signal provided bythe crystal oscillator circuit.

Linear buffer amplifier 720 provides signal isolation, through highinput impedance, as well as amplification of the oscillating (10 MHz)signal produced by the crystal resonator/oscillator driver combination.Linear buffer amplifier 720 is configured to output differential modesignals characterized by linear amplification of the input differentialsignals, that may then be used to drive one or more additional waveshaping-type devices, such as nonlinear buffer amplifiers 1352, 1354 and1356.

In the exemplary embodiment of FIG. 13, the nonlinear buffers 1352, 1354and 1356 function in order to provide signal translation (wave shaping)from the differential sign wave periodic signal present at the output ofthe linear buffer 720 to a digital pulse train at characteristic logiclevels suitable for driving fall-on digital circuit blocks 1358, 1360and 1362. In addition to its signal translation function, nonlinearbuffers 1352, 1354 and 1356 also provide a measure of signalconditioning, transforming the purely sinusoidal signal at their inputsto a very low jittergetter square wave output.

Following digital circuitry 1358, 1360 and 1362 illustrated in theexemplary embodiment of FIG. 13 might be any type of digital circuitrythat requires a stable periodic clock, such as a phase-lock-loop, atunable filter, a digital frequency synthesizer, and the like.Characteristically, high speed switching circuits of these typesgenerate a great deal of noise, particularly as a result of groundbounce, switch transients and ringing. In order to minimize feed throughcoupling of these noise sources back to the crystal oscillator circuit,and in contrast to the prior art, the system of the present inventionutilizes two stages of buffering.

In the prior art, signal transformation from a sinusoidal signal to asquare wave output is typically implemented by using an inverter tosquare sinusoidal input signal. A digital inverter function might becharacterized as a nonlinear amplifier of a transformed sinusoidal inputsignal to a square wave by providing an extremely high gain, such thatthe input signal is driven to the rail during amplification (i.e.,clipping). Thus, the output signal of a typical inverter might becharacterized as a clipped sine wave. This particular nonlinearitycharacteristic of the inverter further provides opportunities for phasenoise to be added to the output signal.

Phase noise (phase jitter) can also be introduced when the slope of asignal waveform going through a zero transition is not sharp. Thus, inthe present invention, phase noise is minimized in the nonlinear bufferamplifiers 1352, 1354 and 1356 by amplifying the differential signalprovided by the crystal oscillator circuit through the linear amplifier720 in order to increase the amplitude, and thus the slew rate, of thesignal prior to its conversion to a square wave. Phase noise resultingfrom zero crossings of the nonlinear buffer amplifiers is therebyminimized.

Further, in a very large scale integrated circuit, there are a greatnumber of digital logic elements coupled to a common power supply.Switching of these digital logic elements causes the power supplyvoltage to move up and down, causing digital switching noise. Thismovement in the power supply induces a jitter component at each inverterthat is used as a buffer in a conventional oscillator circuit. Accordingto the present invention, maintaining a differential signal throughoutthe oscillator circuit, including the wave shaping buffers, allows theeffects of power supply noise to be substantially eliminated from theoscillator, thus maintaining signal quality. In addition, the use of adifferential signal throughout the oscillator's architecture allowscommon mode noise radiated onto the pins of the crystal resonator to berejected.

The number of nonlinear buffers which might be cascaded in order toproduce a suitable clock signal is an additional important feature inthe design of a low phase noise oscillator circuit. In conventionaloscillator circuits, multiple cascaded invertors are used to providehigh isolation of the final, squared output signal. In such cases, eachtime the signal passes through a nonlinear inverter, zero crossingoccurs which offers an additional opportunity for phase noise to beadded to the circuit. In order to minimize phase noise, the presentinvention contemplates a single stage of nonlinear buffering whichpresents a high input impedance to the linear buffer 720 which proceedsit. Additionally, the linear buffer 720 is further provided with a highinput impedance to further isolate the crystal resonator and itsassociated differential oscillator driver circuitry from noise loading.

An exemplary embodiment of a linear buffer suitable for use inconnection with the periodic signal generation circuit of FIG. 13 isillustrated in simplified, semi-schematic form in FIG. 14. The exemplaryembodiment of FIG. 14 illustrates the conceptual implementation of adifferential-in differential-out amplifier. The differentialimplementation has several advantages when considered in practicalapplications. In particular, maximum signal swing is improved by afactor of 2 because of the differential configuration. Additionally,because the signal path is balanced, signals injected due to powersupply variation and switch transient noise are greatly reduced.

The exemplary implementation of a differential-in, differential-outamplifier (indicated generally at 720) of FIG. 14 uses a folded cascadeconfiguration to produce a differential output signal, denoted V_(out).Since the common-mode output signal of amplifiers having a differentialoutput can often be indeterminate, and thus cause the amplifier to driftfrom the region where high gain is achieved, it is desirable to providesome form of common-mode feedback in order to stabilize the common-modeoutput signal. In the embodiment of FIG. 14, the common-mode outputsignal is sampled, at each of the terminals comprising the outputV_(out) and fed back to the current-sink loads of the folded cascade.

Differential input signals V_(in) are provided to the control terminalsof a differential input pair 1464 and 1466, themselves coupled betweenrespective current sources 1468 and 1470 and to a common current-sinkload 1472 to V_(ss). Two additional transistors (P-channel transistorsin the exemplary embodiment of FIG. 14) define the cascade elements forcurrent-sources 1468 and 1470 and provide bias current to the amplifiercircuit.

High impedance current-sink loads at the output of the amplifier 1476and 1478 might be implemented by cascoded current sink transistors(N-channel transistors for example) resulting in an output impedance inthe region of about 1 Mohm. The common mode feedback circuit 1480 mightbe implemented as an N-channel differential pair, biased in their activeregions and which sample the common-mode output signal and feedback acorrecting, common-mode signal into the source terminals of the cascodedtransistors forming the current-sinks 1476 and 1478. The cascade devicesamplify this compensating signal in order to restore the common-modeoutput voltage to its original level.

It should be noted that the exemplary linear amplifier of FIG. 14 mightbe implemented as any one of a number of appropriate alternativeamplifiers. For example, it need not be implemented as a fullydifferential folded cascade amplifier, but might rather be implementedas a differential-in, differential-out op amp using two differential-insingle-ended out op amps. Further, the actual circuit implementationmight certainly vary depending on the particular choices and prejudicesof an analog integrated circuit designer. The input differential pairmight be either an N-channel or a P-channel pair, MOS devices might beused differentially as active resistors or alternatively, passiveresistor components might be provided, and the like. All that isrequired is that the linear amplifier 720 amplifies a differential inputsignal to produce a differential, sinusoidal signal at its output. Thus,the only frequency components reflected back through the linearamplifier 720 will be sinusoidal in nature and thus, will not affect theoperational parameters of the differential crystal oscillator frequency.Further, the linear buffer 720 will necessarily have a relatively highoutput impedance in order to attenuate noise that might be reflectedback from the square wave output of the following nonlinear amplifierstages.

Turning now to FIG. 15, there is depicted a simplified semi-schematicdiagram of a nonlinear buffer, indicated generally at 1582, such asmight be implemented as a wave shaping or squaring circuit 1352, 1354 or1356 of FIG. 13. The nonlinear buffer 1582 receives a differential,sinusoidal input signal at the gate terminals of an input differentialtransistor pair 1584 and 1586. Drain terminals of the differential pair1584 and 1586 are connected together in common and to a current sinksupply 1588 which is coupled to a negative potential. Each of thedifferential pairs' respective source terminals are coupled to a biasnetwork, including a pair of differential bias transistors 1590 and 1592having their gate terminals tied together in common and coupled to aparallel connected bias network. The bias network is suitablyconstructed of a resistor 1594 and a current sink 1596 connected inseries between a positive voltage potential such as Vdd and Vss. A biasnode between the resistor 1594 and current sink 1596 is coupled to thecommon gate terminals of the bias transistor network 1590 and 1592 anddefines a bias voltage for the bias network which will be understood tobe the positive supply value minus the IR drop across bias resistor1594. The current promoting the IR drop across the bias resistor 1594is, necessarily, the current I developed by the current sink 1596.

A differential, square wave-type output (Vout) is developed at twooutput nodes disposed between the respective source terminals of thebias network transistors 1590 and 1592 and a respective pair of pull-upresistors 1598 and 1599 coupled, in turn, to the positive supplypotential. It should be noted, that the bias network, includingtransistors 1590 and 1592, function to control the non-linearamplifier's common mode response in a manner similar to the linearamplifier's common mode network (transistors 1244 and 1246 and resistors1248 and 1250 of FIG. 12).

Although depicted and constructed so as to generate a differentialsquare wave-type output in response to a differential sinusoidal inputsignal, the non-linear buffer 1582 of FIG. 15 is well suited forsingle-ended applications as well as for differential applications. If asingle-ended output is desired, one need only take a signal from one ofthe two symmetric outputs. The choice of whether to implement thenon-linear buffer as a single-ended or a differential buffer will dependsolely on the input requirements of any follow-on digital circuitrywhich the periodic signal generation circuit in accordance with theinvention is intended to clock. This option is solely at the discretionof the system designer and has no particular bearing on practice ofprinciples of the invention.

FIG. 16 is a semi-schematic illustration of an alternative embodiment ofthe differential oscillator driver circuit (718 of FIG. 12). From theexemplary embodiment of FIG. 16, it can be understood that theoscillator driver circuit is constructed in a manner substantiallysimilar to the exemplary embodiment of FIG. 12, except that a crystalresonator is coupled across the circuit halves above the differentialtransistor pair, as opposed to being coupled across a circuit from theVon to Vop output terminals. The alternative configuration of FIG. 16operates in substantially the same manner as the embodiment of FIG. 12and produces the same benefits as the earlier disclosed oscillator. Itis offered here as an alternative embodiment only for purposes ofcompleteness and to illustrate that the specific arrangement of theembodiment of FIG. 12 need not be followed with slavish precision.

It should be understood that oscillator circuits with low phase noiseare highly desirable in many particular applications. FIG. 17illustrates one such application as a reference signal generator in aphase-lock-loop. The phase-lock-loop uses a low phase noise periodicsignal generation circuit in accordance with the invention in order togenerate a reference signal for use by a phase detector. Providing aclean reference signal to the phase detector is fundamental to providinga clean RF output from the PLL. Since noise and nonlinearities inducedby signal generation circuit are carried through the PLL circuit, thusdegrading the RF output, reducing phase noise and providing noiserejection early on in the signal processing chain is advantageous tomaintaining a clean RF output. A differential crystal oscillator (710 ofFIG. 7) advantageously provides this claim signal by maintaining adifferential signal across the terminals of the resonating crystal, animprovement not currently available in state-of-the-art crystaloscillators. Additionally, the use of linear buffer amplifiers followedby nonlinear amplification in a reference oscillator circuit is a uniqueimprovement over the prior art in reducing phase noise.

Since PLLs have become available in integrated circuit form, they havebeen found to be useful in many applications. Certain examples ofadvantageous application of phase-lock-loop technology include trackingfilters, FSK decoders, FM stereo decoders, FM demodulators, frequencysynthesizers and frequency multipliers and dividers. PLLs are usedextensively for the generation of local oscillator frequencies in TV andradio tuners. The attractiveness of the PLL lies in the fact that it maybe used to generate signals which are phase-locked to a crystalreference and which exhibit the same stability as the crystal reference.In addition, a PLL is able to act as a narrow band filter, i.e.,tracking a signal whose frequency may be varying.

A PLL uses a frequency reference source in the control loop in order tocontrol the frequency and phase of a voltage control oscillator (VCO) inthe loop. The VCO frequency may be the same as the reference frequencyor may be a multiple of the reference frequency. With a programmabledivider inserted into the loop, a VCO is able to generate a multiple ofthe input frequency with a precise phase relationship between areference frequency and an RF output. In order to maintain such aprecise phase and frequency relationship, the frequency referenceprovided to the PLL must, necessarily, also be precise and stable.

FIG. 18 is a simplified block diagram of an illustrative frequencysynthesizer that might incorporate the differential periodic signalgeneration circuit of the invention. The frequency synthesizer is asignal generator that can be switched to output any one of a discreteset of frequencies and whose frequency stability is derived from acrystal oscillator circuit.

Frequency synthesizers might be chosen over other forms of frequencysources when the design goal is to produce a pure frequency that isrelatively free of spurious outputs. Particular design goals infrequency synthesizer design might include suppression of unwantedfrequencies and the suppression of noise in a region close to theresonant frequency of the crystal that is a typical source of unwantedphase modulation. Synonymous terms for this type of noise are broadbandphase noise, spectral density distribution of phase noise, residual FM,and short term fractional frequency deviation.

To reduce the noise produced in a synthesizer, crystal oscillators arecommonly used due to their stability and low noise output. The use of aperiodic signal generation circuit incorporating a differential crystaloscillator according to an embodiment of the present inventionadvantageously improves these performance parameters. Improved phasenoise is achieved through the use of linear buffering followed bynonlinear amplification, while noise rejection is provided by thedifferential design utilized throughout the circuitry architecture.

It should be evident that a periodic signal generation circuit accordingto the invention has many uses in modern, state-of-the-art timingcircuits and systems. The periodic signal generation circuit isconstructed of simple active and passive components which are easilyimplemented in modern integrated circuit technology. Thus allowingsubstantially all of the components to be accommodated on one monolithicintegrated circuit chip for which the crystal oscillator portion isproviding a suitable, stable periodic timing reference signal. Only theresonating crystal portion (crystal resonator or quartz crystalresonator) is provided as an off-chip component. This particularconfiguration allows for considerable savings in component parts costsby partitioning more and more functionality into the integrated circuitchip itself.

A more detailed description of the oscillator is provided in U.S. patentapplication Ser. No. 09/438,689 filed Nov. 12, 1999 (B600:33758)entitled “Differential Crystal Oscillator” by Christopher M. Ward andPieter Vorenkamp; based on U.S. Provisional Application No. 60/108,209filed Nov. 12, 1998 (B600:33588), the subject matter of which isincorporated in its entirety by reference. The oscillator's output is adifferential signal that exhibits high common mode noise rejection. Useof a low noise reference oscillator with differential signaltransmission allows the synthesis of stable low noise local oscillatorsignals. Advantageously in the present exemplary embodiment of theinvention a unique generation of the local oscillator signals allowscomplete integration of a receiver circuit on a CMOS integrated circuitby reducing noise in the signal path.

Frequency synthesizers and a radio frequency receiver often incorporatephase locked loops that make use of a crystal oscillator as a frequencyreference. A PLL is used to provide coarse tuning within the frequencyband of interest while a second PLL provides fine tuning steps.Advantageously, the present embodiments of the invention utilize amethod of coarse/fine PLL adjustment to improve the performance of theintegrated tuner.

Coarse/Fine PLL Adjustment

FIG. 19 is a diagram illustrating receiver tuning. The combination of awide band PLL 1908 and a narrow band PLL 1910 tuning provides acapability to fine tune a receiver's LOS 1902, 1904 over a largebandwidth in small frequency steps. For the exemplary embodiments of QAMmodulation a small frequency step is 100 kHz, and 25 kHz for NTSCmodulation. Fine tuning is available over an entire exemplary 50 MHz to860 MHz impact frequency band width 1906. The first PLL 1908 tunes afirst LO 1902 in large 10 MHz frequency steps and the second PLL 1910tunes a second LO 1904 in much smaller steps. The first intermediatefrequency (IF) filter 1912 has a sufficiently wide band width to allowup to 10 MHz frequency error in tuning the first intermediate frequency,with the narrow band PLL providing final fine frequency tuning toachieve the desired final IF frequency 1914.

FIG. 20 is a block diagram of an exemplary tuner 2002 designed toreceive a 50 to 860 MHz bandwidth signal 2004 containing a multiplicityof channels. In this exemplary band of frequencies, there are 136channels with a spacing between channel center frequencies of sixmegahertz 2008. The tuner selects one of these 136 channels 2006 thatare at a frequency between 50 and 860 MHz by tuning to the centerfrequency of the selected channel 2010. Once a channel is selected thereceiver rejects the other channels and distortion presented to it. Theselected channel is down converted to produce a channel centered about a44 MHz intermediate frequency (IF) 2012. Alternatively the value of theintermediate frequency ultimately produced by the tuner may be selectedutilizing the method of the invention to provide any suitable final IFfrequency, such as 36 MHz.

In selecting one of these 136 channels, a maximum frequency error in thelocal oscillator (LO) frequency used to tune the channel to a given IFof plus or minus 50 kHz is allowable. Using one frequency conversion todirectly tune any one of the 136 channels to 44 MHz would require atuning range in the local oscillator of 810 MHz. This would require alocal oscillator that tunes from 94 to 854 MHz, if utilizing high sideconversion.

Achieving this with a single LO is impractical. Tuning range in localoscillators is provided by varactor diodes that typically require 33volts to tune them across their tuning range. Additionally, within thistuning range a frequency tuning step of 100 kHz is required to ensurethat the center frequency of a tuned channel is tuned within plus orminus 50 kHz. Thus, a large range of frequencies would have to be tunedin small increments over a 33 volt tuning signal range.

Returning to FIG. 19 illustrating the frequency tuning method of theinvention an exemplary 50 to 860 MHz signal 1906 is presented to a firstmixer 1916 that is tuned with a wide band PLL 1908 that tunes a first LO1902 in frequency steps of 10 MHz. This local oscillator 1902 is set toa frequency that will nominally center a channels that has been selectedat a first IF of 1,200 MHz 1918. The first IF 1918 is then mixed 1920 tothe second IF of 275 MHz 1922. This is done by the narrow band PLL 1910that tunes a second LO 1904 in frequency steps of 100 kHz. The second IF1922 is next mixed 1924 down to a third IF 1926 of 44 MHz by a thirdlocal oscillator signal 1928. This third local oscillator signal 1930 isderived from the second local oscillator or narrow band PLL signal bydividing its frequency by a factor of four.

FIG. 21 is an exemplary table of frequencies utilizing coarse and finePLL tuning to derive a 44 MHz IF (“IF-3”). A process is utilized todetermine the wide and narrow band PLL frequencies. The relationshipbetween the wideband PLL and narrowband PLL frequencies to yield thedesired intermediate frequency is found from:

FLO1−Fsig−(5/4*FLO2)=Fif   (4)

where:

FLO1: PLL1 frequency (10 MHz steps)

FLO2: PLL2 frequency (e.g., 25 kHz/100 kHz/200 kHz or 400 kHz step)

Fsig: Input signal

Fif (e.g., 44 MHz or 36 MHz or whatever IF is required)

Example:

1250M−50M−(5/4*924.8M)=44M

where:

Fsig=50 MHz

FLO1=1250 MHz

FLO2=924.8 MHz

Fif=44 MHz

FIGS. 21 and 22 utilized this formula to derive the values entered intothem to tune the exemplary cable TV signals “Frf”. For example the firstcolumn 2102 of the table lists the frequencies needed to tune a signalcentered at 50 MHz (“Frf”) to a 44 MHz final IF (“IF-3”). To tune areceived channel centered at 50 MHz a first LO of 1,250 MHz (“LO-1”) isprovided by a wide band, or coarse, PLL. This produces a first IF of1,200 MHz (“IF-1”). Next utilizing 100 kHz tuning steps to adjust LO 2,it is set to 924.8 MHz (“LO-2”). Note this is not exactly 925 MHz.Dividing the second LO by 4 in this instance yields 231.2 MHz for athird LO (“LO-3”). When LO 3 is applied to the second IF of 275.2 athird IF of 44 MHz (“IF-3”) is produced. This tuning arrangement isillustrated for received channels having a six MHz channel spacing ascan be seen from the line entitled “Frf”. In each case the coarse finetuning approach yields a third IF (“IF-3”) of 44 MHz.

FIG. 22 is an illustration of an alternative embodiment of the coarseand fine PLL tuning method to produce an exemplary final IF of 36 MHz.In this case as previously, a first IF (IF-1)is tuned to 1,200 MHz plusor minus 4 MHz. And second LO (LO-2) is close to 930 MHz, utilizing asmall offset to yield a third IF of 36 MHz (IF-3). These predeterminedtuning frequencies are stored in a memory and applied when a command isgiven to tune a given channel. Alternatively an algorithm may besupplied to produce the tuning frequencies. It is understood that thesefrequencies are exemplary and other frequencies that utilize this methodare possible.

Thus, it can be seen that the interaction of course and fine PLLfrequencies are utilized to produce a third IF of 44 MHz. A second LO(LO-2) is maintained close to a frequency of 925 MHz to tune each of thechannels. However, it is slightly off by a very small tuning step of 100kHz. Note that the first IF (IF-1) is not always right at 1,200 MHz.Sometime it is off by as much as 4 MHz either above or below 1,200 MHz.This error will still result in signal transmission through a first IFfilter. The maximum error utilizing this scheme is plus or minus 4 MHz.

This method of PLL adjustment is described in more detail in U.S. patentapplication Ser. No. 09/438,688 filed Nov. 12, 1999, (B600:34015)entitled “System and Method for Coarse/Fine PLL Adjustments” by PieterVorenkamp, Klaas Bult and Frank Carr; based on U.S. ProvisionalApplication No. 60/108,459 filed Nov. 12, 1998 (B600:33586), the subjectmatter of which is incorporated in its entirety by reference.

A coarse, and a fine PLL use a common reference frequency oscillator.Local oscillator signals produced by the frequency synthesizer's phaselocked loops inject noise produced in the reference frequency oscillatorand the PLLs into a signal path through the PLL output. Noise injectedcan be characterized as either phase noise or jitter. Phase noise is thefrequency domain representation of noise that, in the time domain ischaracterized as jitter. Phase noise is typically specified as a powerlevel below the carrier per Hertz at a given frequency away from thecarrier. Phase noise can be mathematically transformed to approximate ajitter at a given frequency for a time domain signal. In a clock signaljitter refers to the uncertainty in the time length between zerocrossings of the clock signal. It is desirable to minimize the jitterproduced in an oscillator circuit and transmitted through the signalchain into the signal path to prevent noise degradation in the receiverpath. Equivalently, any oscillator producing a stable output frequencywill suffice to produce a reference frequency for the PLL circuitry.

Another obstacle to integrating an entire receiver on a single CMOS chiphas been the inability to fabricate a satisfactory filter structure onthe chip. As previously described, a multitude of unwanted frequenciescreated through circuit non linearities are a major obstacle inachieving satisfactory receiver performance. Filtering is one method ofeliminating these unwanted spurious signals. An integrated filter'scenter frequency tends to drift, and needs calibration to maintainperformance. To successfully use filtering on chip, an auto calibrationloop is needed to center the filter response.

FIG. 23 is a block diagram of a dummy component used to model anoperative component on an integrated circuit chips. According to oneaspect of the invention, a dummy circuit on an integrated circuit chipis used to model an operative circuit that lies in a main, e.g. RF,signal path on the chip. Adjustments are made to the dummy circuit in acontrol signal path outside the main signal path. Once the dummy circuithas been adjusted, its state is transferred to the operative circuit inthe main signal path. Specifically, as shown in FIG. 23, there is a mainsignal path 2201 and a control signal path 2202 on an integrated circuitchip. In main signal path 2201, a signal source 2203 is coupled by anoperative circuit 2204 to be adjusted to a load 2205. Main signal path2201 carries RF signals. Signal source 2203 generally represents theportion of the integrated circuit chip upstream of operative circuit2204 and load 2205 generally represents the portion of the integratedcircuit chip downstream of operative circuit 2204. In control signalpath 2202, a control circuit 2206 is connected to a dummy circuit 2207and to operative circuit 2204. Dummy circuit 2207 is connected tocontrol circuit 2206 to establish a feedback loop. Dummy circuit 2207replicates operative circuit 2204 in the main signal path in the sensethat, having been formed in the same integrated circuit process asoperative circuit 2204, its parameters, e.g., capacitance, inductance,resistance, are equal to or related to the parameters of operativecircuit 2204. To adjust operative circuit 2204, a signal is applied bycontrol circuit 2206 to dummy circuit 2207. The feedback loop formed bycontrol circuit 2206 and dummy circuit 2207 adjusts dummy circuit 2207until it meets a prescribed criterion. By means of the open loopconnection between control circuit 2206 and operative circuit 2204 thestate of dummy circuit 2207 is also transferred to operative circuit2204, either on a one-to-one or a scaled basis. Thus, operative circuit2204 is indirectly adjusted to satisfy the prescribed criterion, withouthaving to be switched out of the main signal path and without causingdisruptions or perturbations in the main signal path.

In one implementation of this dummy circuit technique described below inconnection with FIGS. 24a-c and FIGS. 25-27, operative circuit 2204 tobe adjusted is a bank of capacitors in one or more operative bandpassfilters in an RF signal path, dummy circuit 2207 is a bank of relatedcapacitors in a dummy bandpass filter, and control circuit 2206 is aphase detector and an on-chip local oscillator to which the operativefilter is to be tuned. The output of the local oscillator is coupled tothe dummy filter. The output of the dummy filter and the output of thelocal oscillator are coupled to the inputs of the phase detector tosense the difference between the frequency of the local oscillator andthe frequency to which the dummy filter is tuned. The output of thephase detector is coupled to the dummy filter to adjust its bank ofcapacitors so as to tune the dummy filter to the local oscillatorfrequency. After the dummy filter is tuned, the state of its capacitorbank is transferred, either on a one-to-one or scaled basis, to theoperative filter. Since the capacitor bank in the dummy filterreplicates that of the operative filter, the frequency to which theoperative filter is tuned can be easily scaled to the frequency of thedummy filter.

In another implementation of the dummy circuit technique described belowin connection with FIGS. 28 to 33, operative circuit 2204 to be adjustedis a filter having a spiral inductor that has a temperature sensitiveinternal resistance. Dummy circuit 2207 has an identical spiralinductor. Control circuit 2206 has a controllable variable resistor inseries with the inductor of dummy circuit 2207. The controllableresistor is driven by a feedback loop to offset changes in the internalresistance of the inductor of dummy circuit 2207. Operative circuit 2204has a similar controlled resistor in series with its inductor totransfer the resistance value of the controllable resistor in controlcircuit 2206 to the resistor of the operative circuit 2204 in open loopfashion.

Filter Tuning

FIG. 24a is a block diagram illustrating the use of a tuning circuitoutside of a signal path to tune bandpass filters present in a receiver.A tuning circuit 2302 utilizes a substitute or “dummy” filter stage 2310to derive tuning parameters for a filter bank 2304 present in a signalpath 2306. The tuning circuit utilizes a local oscillator signal 2308available in the receiver to tune the dummy filter 2310 to the centerfrequency of the local oscillator. Once tuned, the dummy filters 2310tuned component values that result in a tuned response at the localoscillator frequency are scaled in frequency and applied to the bandpassfilter 2312. The filters are tuned at startup, and the tuning circuitryis turned off during normal operation. This prevents the injection ofadditional noise into the signal path during operation.

FIG. 24b is a flow diagram of the tuning process in operation receiveris initially powered up 2312 and local oscillator signals generated byPLLs are centered at their design frequency 2314. Once the PLLs arelocked their frequency is a known condition. Next substitute filtertuning is initiated 2316 and performed. When finished a signal isreceived back from the filter tuning network indicating that it is ready2318. Information from the tuning network is copied to the receive pathfilter circuit 2320. Next the filter tuning circuit is turned off 2322disconnecting it from the filter circuit. In the embodiments of theinvention the narrow band PLL (2308, of FIG. 24a) is used as referencefrequency in the tuning circuit. However, it is understood that thistuning technique may be utilized with any readily available signal.

Returning to FIG. 24a, in an exemplary embodiment of the invention a 925MHz signal is directly available from the narrow band PLL 2308. It isused to tune the dummy filter 2310 contained in the tuning circuit 2302associated with the 1,200 MHz filter 2304. After the dummy filter istuned to 925 MHz, frequency scaling is used to obtain the propercomponent values for the 1,200 MHz filter response to be centered. Theexemplary 925 MHz signal generated by the narrow band PLL is divided by4 to yield a 231 MHz third LO signal utilized in additional tuningcircuitry.

Other divisions or multiplications may be equivalently used to tunedummy filters. A second exemplary filter tuning circuit 2302 for a 275MHz filter contains a dummy filter 2310 that is tuned to a centerfrequency of 231 MHz. Once tuned, the component values used to centerthe 231 MHz dummy filter 2310 are scaled to yield a centered responsefor the 275 MHz filter 2304. At this point in time the tuning circuits2302 are switched off. It is especially important to turn off theexemplary tuning circuits on the 275 MHz filter since the 231 MHz signalused to tune its dummy filter falls in an exemplary 50-860 MHz band.

It is to be understood that any available frequency may be used to tunea substitute filter so that another filter, that does not have anappropriate tuning signal present, may be tuned. This is done by scalingthe component values of the tuned dummy filter to values appropriate forthe filter not having the tuning frequency present. Tuning valuesobtained for a dummy filter may be applied to all filters present in abank of filters having a common center frequency. Also tuning valuesobtained for a dummy filter may be applied to multiple filters presenthaving differing center frequencies by applying differing scalingfactors. Finally multiple filters at different locations in a signalpath that have common center frequencies may be tuned by a common tuningcircuit.

Capacitors disposed on an integrated circuit vary in capacitance valueby as much as +/−20%. Thus, to provide a satisfactory receiverperformance a method of tuning integrated filters that removes thisvariation in capacitance is needed. In an LC filter circuit either aninductance or a capacitance can be tuned. However, inductors aredifficult to tune. Therefore, in the embodiments of the invention valuesof capacitance present in the filters are tuned. In tuning the exemplaryembodiments, one or more capacitors are switched in and out of an LCfilter circuit to tune it.

These capacitors are switched in and out of a filter circuitelectronically. Capacitors with the same dimensions are provided in abandpass filter and a dummy filter to provide satisfactory matchingbetween the devices. Switchable caps in the embodiments of the inventionare MOS caps that are all of the same value and from factor. However, itis to be recognized that other weighting of capacitor values could beprovided to achieve an equivalent function. For example, binary or 1/×weighted values of capacitors could be disposed in each filter toprovide tuning. In the embodiments of the invention a bank of fixedcapacitors and a bank of electronically tunable capacitors are provided.The adjustable capacitors in the exemplary embodiment represent 40% ofthe total capacitance provided. This is done to provide for the ±20%variance in center frequency due to manufacturing variances. Toaccommodate other ranges of manufacturing variations or alternativetuning schemes any fraction or all of the capacitors may be switchable.It is also understood that any type of switchable capacitor, in additionto a MOS capacitor type may be utilized.

FIGS. 24a-24 c are exemplary illustrations of a tuning process utilizingswitched capacitors. Filter responses shown at the bottom plot 2402illustrate a tuning of a dummy filter 2310 that is contained in a tuningcircuit 2302 of FIG. 24a. A frequency response being tuned in the uppergraph 2404 shows the tuning of the exemplary 1,200 MHz bandpass filter2304 of FIG. 24a. Initially none of the switched capacitors are appliedin a dummy filter circuit. This places the filter response initially2406 above the final desired tuned response frequency 2408. In thisexample capacitors are added until the filter response of the dummyfilter is centered about 925 MHz. However, the tuned response of the 925MHz dummy filter 2408 is not the desired center frequency of thebandpass filter in the signal path. The values used in to tune the dummyfilter would not tune the 1,200 MHz filter to the correct response.Frequency scaling is used to tune the desired response. This can beachieved because identical capacitors disposed on a chip are very wellmatched in value and parasitics. In particular capacitor matching iseasy to achieve by maintaining similar dimensions between groups ofcapacitors. In scaling a response to determine a capacitance to apply ina bandpass filter, identical inductance values have been maintained inthe dummy and bandpass circuits. Thus, only a scaling of the capacitorsis necessary. The frequency relation in the exemplary embodiment isgiven by the ratio: $\begin{matrix}{\frac{1}{2} \approx \sqrt{\frac{( L_{2} )( C_{2} }{( L_{1} )( C_{1} }}} & (5)\end{matrix}$

For this particular embodiment utilizing identical inductor valuesL₁=L₂. This reduces to: $\begin{matrix}{\frac{f_{1}}{f_{2}} \approx \sqrt{\frac{( C_{2} )}{( C_{1} )}}} & (6)\end{matrix}$

For the exemplary embodiment this is equal to 925/1200, or a capacitanceratio of 3:5. However, it is understood that other ratios will allowtuning to be performed equivalently.

Returning to FIG. 24a various control signals applied to the tuningcircuit are shown. In the event that the tuning is slightly off afterthe tuning procedure, an offset control circuit is provided within thetuning circuit of FIG. 24a to move the tuning of the filters up or downslightly by providing a manual means of adding or removing a capacitor.This control is shown by an “up/down” control line 2324 of FIG. 24a. Theexemplary tuning circuit of FIG. 24a is additionally provided with a“LO” 2308 tuning frequency to tune the dummy filter. The “10 MHzreference” signal 2326 is utilized as a clock in the tuning circuit thatcontrols the sequence of adding capacitors. The “reset” signal 2328resets the tuning circuit for the next tuning cycle.

FIG. 25 is a block diagram of an exemplary tuning circuit. A resetsignal 2502 is utilized to eliminate all the capacitors from the circuitat power up by resetting a counter 2504 that controls the application ofthe switched capacitors. The reset signal may be initiated by acontroller or generated locally. This provides a known starting pointfor filter tuning. Next a filter figure of merit is examined todetermine iteratively when to stop tuning.

FIG. 26 illustrates the amplitude 2602 and phase 2604 relationship in anLC filter tuned to its center frequency, fc. In tuning a filter to acenter frequency two responses are available for examination. Amplitudeand phase response are parameters that may be used to tune the filter.For a wide band LC filter amplitude response 2602 is not the optimalparameter to monitor. At the center frequency the top of the responsecurve is flat making it difficult to judge if the response is exactlycentered. The phase response 2604 however, has a rather pronounced slopeat the center frequency. The steep slope of the phase signal provides aneasily discernable transition for determining when the center frequencyhas been reached.

Returning to FIG. 25, phase detection is used to detect when a dummyfilter 2506 has been tuned. An exemplary 925 MHz input from a narrowband PLL is input 2508 to a phase detector 2510. The phase detectorcompares the phase of a signal input to a dummy filter 2508 to a phaseof the output 2512 of that filter 2506. The phase detector produces asignal that is internally low pass filtered to produce a DC signal 2514proportional to the phase difference of the two input signals 2512,2508. When tuned there is a 90 degree phase shift across capacitorsinternal to the phase detector, that corresponds to 0 degrees of phaseshift across the filter. Zero (0) degrees of phase shift produces a 0volt output. Since it is known that with the capacitors switched out ofthe filter circuit 2506 that the center frequency of the filter is high,the comparator 2516 following the low pass filter is designed to output2518 a high signal that enables filter capacitors to be switched inuntil the phase detector 2510 indicates no phase difference is presentacross the filter 2506 at the tuned frequency. With a zero degree phaseshift detected the comparator 2516 disables the counter preventing anyfurther capacitors from being switched into the filter circuit.

The phase detector 2510 of the exemplary embodiment utilizes a gilbertcell mixer 2512 and an integral low pan filter 2525 to detect phase.However, other phase detectors may be equivalently substituted for themixer circuit. The 90° phase shift between an i port 2508 and a q port2512 is being detected by the mixer. A 90° phase shift between the i andthe q signals in the mixer provides a 0 volt output indicating thatthose signals are in quadrature relation to each other. The signals areshown as differential signals, however single ended signals mayequivalently be used.

The phase detector out 2514 is next fed into a comparator 2516 that isset to trip on a zero crossing detected at its input. When a zerocrossing is encountered as the phase detector output approaches zero,the comparator latches and a counter 2504 is shut off and reset 2518.The comparator function is equivalently provided by any standardcomparator circuit known by those skilled in the art.

The counter 2504 counts based on the 10 MHz reference clock 2524,although many periodic signals will suffice as a clock. As the counteradvances more filter capacitors are switched into the circuit. In theembodiments of the invention 15 control lines 2526 are used tosimultaneously switch the capacitors into the dummy filter and thebandpass filter bank. The control lines remain hard wired to bothfilters 2528, 2506, and are not switched off. However, once thecomparator 2516 shuts the counter 2504 off the tuning circuit 2530 isinactive and does not affect the band pass filter 2520 in the signalpath.

FIG. 27 is a schematic diagram showing the internal configuration ofswitchable capacitors in a differential signal transmission embodimentof the dummy filter 2506 and the construction of the phase detector2510. A set of fifteen control lines 2526 are utilized to switch fifteenpair of MOS capacitors 2702 on and off. The capacitors are switched inand out by applying a given control signal to a virtual ground point2704 in this configuration. Thus, when the capacitors are connected asshown the control signal is being applied at a virtual ground. Thus,parasitic capacitances at this point will not affect the filter 2506performance. A gain producing LC stage 2706 of the dummy filter is of adifferential configuration and has its LC elements 2708 connected inparallel with the MOS capacitors 2702.

Thus, with a capacitance ratio of 3:5 being utilized in the exemplaryone line of embodiment a hard wired bus 2526 going to the dummy filter2506 will switch in 5 unit capacitors, while the other end of the linethat goes to the bandpass filter (2528 of FIG. 25) in the signal pathwill switch in 3 unit capacitors.

In the mixer circuit that is used as a phase detector 2710 in theexemplary embodiment, differential image (“i”) signals I_(P) and I_(N)and differential quadrature (“q”) signals Q_(P) and Q_(N) are input tothe phase detector. A conventional Gilbert cell mixer configured as aphase detector 2710, as shown, has delay between the i port 2508 and qport 2512 to the output 2514. The i delay to the output tends to belonger due to the fact that it must travel through a greater number oftransistors than the q input to output path. Thus, even if i and q areexactly 90 degrees out of phase a DC offset tends to produced due to thepath length differences causing a phase error. To remedy this situationa second Gilbert cell mixer is duplicated 2710 and connected in parallelwith the first 2710. However, the i port and the q port connected to themixer 2712 are swapped to average out the delay thus tending to reducethe offset. This results in an almost 0° output phase error that isindependent of frequency. Other types of phase detectors and other meansof equalizing the delay, such as a delay line are understood by thoseskilled in the art to provide an equivalent function.

In the embodiment shown, the loss pass filter is implemented by a singlecapacitor 2714 at each output. However, other equivalent methods ofachieving a low pass filter known to those skilled in the art areacceptable as well.

A method of filter tuning the advantageously uses the frequencysynthesizer output is fully described in U.S. patent application Ser.No. 09/438,234 filed Nov. 12, 1999 (B600:34013) entitled “System andMethod for On-Chip Filter Tuning” by Pieter Vorenkamp, Klaas Bult andFrank Carr; based on U.S. Provisional Application No. 60/108,459 filedNov. 12, 1998 (B600:33586), the subject matter of which is incorporatedin its entirety by reference.

Filters contain circuit elements whose values are frequency andtemperature dependent. The lower the frequency, the larger the size ofthe element required to realize a given value. These frequency dependentcircuit elements are capacitors and inductors. The fabrication ofcapacitors is not as problematic as the fabrication of inductors on anintegrated circuit. Inductors require relatively more space, and becauseof their size has a temperature dependent Q.

Active Filter Multi-Track Integrated Spiral Inductor

FIG. 28a is a plan view of a multi-track spiral inductor 2800 suitablefor integration onto an integrated circuit, such as one produced with aCMOS process. A standard CMOS process often utilizes a limited number oflayers and a doped substrate. These conditions do not provide optimumconditions for fabrication an on chip inductor. Currents induced in theheavily doped substrate tend to be a source of significant losses. Themulti-track inductor 2800 is made from several long narrow strips ofmetal 2804, 2806 connected in parallel 2808, 2810 and disposed upon anintegrated circuit substrate 2802. A multi-track integrated spiralinductor tends to produce an inductance with a higher Q. High Q isdesirable to achieve lower noise floors, lower phase noise inoscillators and when used in filters, a better selectivity. To reduceseries resistance and thus improves the Q of a spiral inductor, a singlewide track width in the spiral is typically used by those skilled in theart.

Skin effect is a frequency dependent phenomena, occurring where a givencurrent is present in a conductor, that produces a current density inthe conductor. At DC, where the frequency is zero, the current densityis evenly distributed across a conductor's cross section. As thefrequency is increased the current crowds to the surface of theconductor. At high frequency substantially all of the current tends toflow in the surface of the conductor. Thus, the current density at thecenter of the conductor is very low, and at the surface it is greater. Askin depth is the depth in the conductor (δ) at which the current is1/e=0.368 the value of the current on the surface. The equation for skindepth is:

δ=(2πfσμ)^(−1/2)  (7)

where:

f=frequency in Hz

σ=conductivity of the conductor in mhos/m

μ=permeability in Henrys/m

As can be seen from the equation (7) the frequency increases the skindepth decreases.

When track width is increased beyond 10-15 μm the skin effect causes theseries resistance of a spiral inductor to increase at high frequencies.Thus, Q is reduced even though a wide track has been used. This trendtends to limit the maximum Q achievable in integrated spiral inductors.

Reduced Q at high frequencies in spiral inductors having a wide trackwidth tends to be caused by eddy currents induced in a spiral inductor'sinner sections 2812. Multiple narrow tracks placed side-by-side 2804,2806 tends to reduce the eddy currents produced. In a spiral inductoreddy currents tend to produce a magnetic field opposing a desiredmagnetic field that produces a desired inductance. Thus, by reducing theeddy currents the desired inductance is more efficiently produced withless loss, hence raising the inductor's Q.

The multi-track technique is advantageously utilized in applicationsrequiring a winding. Examples of devices utilizing multi-track windingscomprise: planar spiral inductors (rectangular, octagonal or circularpatterns) transformers, and baluns. These devices are suitable forincorporation into architectures comprising: integrated circuits, hybridcircuits, and printed circuit boards.

The first exemplary embodiment shown in FIG. 28a is of a square spiralinductor 2800 that is wound in two turns with several narrow tracks2804, 2806 disposed in parallel upon a substrate 2802. Equivalently anynumber of track may be used to achieve a multi-track design. A turn iscounted each time the track is wound around in a spiral such that astarting point 2814 is passed. Typically 5 to 20 turns are utilized in aspiral, with 3 to 10 producing optimum performance. Alternativeembodiments of the invention equivalently utilize one or more turns asrequired to achieve a desired inductance for a given track width.

For example a single track spiral inductor is designed to have a singletrack width of 30 μm in a given number of turns to produce a desiredinductance. By splitting an exemplary 30 μm wide track into two 15 μmtracks 2804, 2806 disposed in parallel on the substrate, the inductor Qtends to increase. A typical Q for the single track inductor with atrack-width of 30 mm is 5.14. The Q of the exemplary dual track inductor2800 with two 15 mm tracks 2804, 2806 in parallel is typically 5.71.Thus, utilizing two narrower tracks in parallel tends to yield animproved Q over a single wider track. A typical improvement in Q forsplitting an inductor's track is in excess of 10%. A further splittingof an inductor's tracks into multiple narrower parallel tracks tends tofurther increase the measured Q.

FIGS. 28b-28 g illustrate various planar devices comprising inductor2820, 2822, 2824, 2816 and transformer 2826, 2818 configurationssuitable for incorporating multiple tracks into their designs. Thedevices are shown with single tracks for clarity. However, it isunderstood that each of the tracks shown in the devices may comprisemultiple tracks constructed as described below. The method isadvantageously used in, various planar inductor topologies comprisingsquare 2820, octagonal 2822, and circular 2824.

An example of a 3-turn symmetric inductor is shown 2816. Each of thesingle tracks shown is sub-divided into multiple tracks as describedbelow. The multiple tracks are joined only at the ends 2826. A series ofphantom lines 2828 indicate tracks on a different layer, connected to atrack shown by a solid line using one or more vias. When routingmultiple vertical tracks 2825 that are tied in common with vias 2827 toa different layer the tracks being routed may be reduced to one track2829, or the multiple vertical structure may be maintained 2831. Thismethod is suitable for suitable for symmetric inductors of any number ofturns.

The symmetric inductor 2816 may be used as a building block to constructa transformer 2818. A second symmetric inductor 2833 is wound inparallel with the symmetric inductor shown 2816. The ends of the firstinductor 2830, 2832 are kept separate from the second symmetric inductor2834, 2836. The resulting four ends 2830, 2832, 2834, 2836 comprise thetransformer connections. The symmetric inductor with a parallel winding2818 is suitable for use as a balun for converting single-ended signalsto differential signals and vice versa. The coupling is provided by thewinding arrangement.

Alternatively two symmetric inductors of the type shown 2816 are placedsubstantially on top of each other, on different layers to produce atransformer, or balun as previously described.

FIG. 28h is an illustration of a second embodiment of an inductor havinga single winding comprising five tracks 2838 per layer. The tracks are amaximum of 5 μm wide. The embodiment comprises one or more layers. Thesecond embodiment further comprises a square spiral form factorconstructed from five conductive tracks 2838 per layer formed into asingle turn. Individual tracks are kept at a maximum width of 5 μm. A0.6 μm gap between adjacent tracks 2840 is maintained. The minimum gapis a requirement for a given process. Here it is a limitation of theCMOS process. At frequencies between 2 GHz and the inductor'sself-resonant frequency an inductor constructed of multiple tracks ofwidths up to the maximum width tends to exhibit improved performance inquality factor (Q). Utilizing multiple narrower tracks in parallel tendsto yield an improved Q over a single wider track, and a single doubletrack inductor. The tracks in each layer are connected at their ends bya conductive strip 2842.

In a third exemplary embodiment six tracks are disposed on a layer. Inthe embodiment, a 30 mm track inductor is split into six parallel tracksof 5 mm each. Utilizing 6 tracks tends to improve the Q from 5.08 to8.25, a 62% increase in Q. Improvements in an inductor's quality factortends to improve the suitability of spiral inductors for use in highfrequency circuits. For example multi-track spiral inductors areadvantageously used in high frequency voltage-controlled oscillator(VCO) and tuned amplifier circuits.

FIG. 28i illustrates the placement of tracks 2844, 2848 in a layeredstructure 2846. In constructing an inductor according to this techniquea set of parallel tracks 2844, 2848 are disposed side-by-side in aarraignment similar to that of coupled transmission lines. The side byside pattern is disposed in multiple layers M5, M4, M3. Each trackdisposed in a common layer has a starting point and an ending point.Each track's starting point 2850 in a layer is coupled together, andeach track's ending point is coupled together in the layer 2852. A passthrough track 2854 is disposed in a layer to provide access to the endof an inner turn.

The placement of conductive via holes V2, V3, V4 in the embodiments ofthe invention couple the tracks in adjacent layers M2, M3, M4, M5. Inthe multiple track inductors described, the multi-tracks are joinedtogether at the beginning of a winding 2850 and again joined together atthe end of the winding 2852 by a conductive material. Vias betweenlayers are formed to couple a bottom track to one or more tracksdisposed in layers above it. Vias are utilized along the length of thetrack.

Thus, by utilizing this technique a group of multiple tracks are formedin a first embodiment by disposing tracks in a combination of verticallayers M2, M3, M4, M5 and side-by-side in the same layer 2856, 2858. Ina second embodiment an inductor is formed by disposing tracks side byside in the same layer. In a third embodiment an inductor is formed bylayering tracks on top of each other vertically. By connecting the tracklayers vertically using vias, the series resistance loss tends to bedecreased due to increased conductor thickness.

For example, in an embodiment three layers are utilized in whichindividual track width is limited to 5 to 6 μm in width, with four tosix tracks disposed in parallel in each layer. In the embodiment viasare used vertically between metal layers to connect the tracks. The viasare used in as many places as possible along the length of each track tocouple the layers. However, the parallel tracks in the same layer arejoined to each other only at the ends.

FIG. 28j is an illustration of an embodiment utilizing a shield 2860disposed beneath an inductor 2862. A shield tends to double inductor Qin the 3˜6 GHz frequency range for a lightly doped substrate, such as isutilized in a non-epi process, a 100% improvement. If a heavily dopedsubstrate, such as is found in an epi-process is utilized, the shieldtends not to improve inductor Q. The embodiment shown utilizes an n⁺shield 2860. An n⁺ diffusion advantageously tends to possess lesscapacitance between the inductor and ground plane than if polysilicon isused as the shield material. The ground planes are silicided n⁺ materialpossessing a low resistivity. Silicided n⁺ material is available in thefabrication process utilized in CMOS.

FIG. 28k is an illustration of a patterned shield 2864 that is utilizedbeneath a multi-track inductor. A patterned n⁺ shield is utilizedbeneath the inductor to reduce losses to the substrate. In theembodiment an n+ diffusion is provided in a fingered pattern of n+regions 2866. Polysilicon is disposed in a series of gaps 2868 betweenthe n⁺ fingers. The patterned shield provides shielding equivalent to asolid ground plane, but without undesirable eddy currents. The shield isdisposed in a fingered pattern 2866 to prevent having a single largesurface as a ground plane. Fingering tends to prevent the inducement ofeddy currents flowing in one or more ground loops. Ground loops tend tocancel the inductance produced in the spiral.

The finger structure of the patterned shield is constructed from an n⁺diffusion layer. The gaps between the fingers are filled withpolysilicon material. The n⁺ diffusion fingers and polysilicon fingersformed by the filling are not coupled to each other, thus preventingeddy current flow in the shield. An interdigitated shield 2864 asdescribed above tends to be an improvement over an n⁺ only shield 2860of FIG. 28j. The interdigitated n⁺ finger shield also tends to be animprovement over a higher capacitance fingered polysilicon shield havinggaps between the fingers, which is known in the art.

The individual fingers of like material are connected 2870. To suppresseddy currents and break ground loops care is taken in the connection ofindividual fingers 2886 in a ground shield pattern. The ends of thefingers in a row are connected by a conductive strip of metal 2870. Thisconnection is repeated at each grouping. The groupings are connected2870 to a single ground point 2874. In an embodiment a ring ofconductive material is disposed on the substrate to connect the fingerpatterns.

A cut 2876 in the ring is added to suppress ground loop currents. Thecut maintains a single point ground by only allowing the flow of currentin one direction to reach the single point ground 2874.

One or more spirals of metal have a series resistance associated withthem. A spiral can be quite long, thus, the series resistance of theinductor is not negligible in the design of the circuit even with aparallel connection of tracks. As the temperature of the circuit rises,such as would occur after the initial power-up of an integrated circuit,the series resistance of the inductor increases, thus causing the Q todecrease. Circuitry is provided to continuously compensate for thisincreasing series resistance.

An inductor, or coil, has always been a fabrication problem inintegrated circuitry. Inductors are typically not used in integratedcircuits due to the difficulty of fabricating these devices with highQ's and due to the large amount of area required to fabricate them.

It is a rule of thumb that the higher the frequency the smaller thedimensions of the integrated circuit component required in a filter toachieve a given set of circuit values. A spiral inductor of the typedescribed in the embodiments of the invention allows an inductor withimproved Q's to be satisfactorily fabricated on a CMOS substrate. Manyalternative embodiments of the spiral are known to those skilled in theart. The realization of inductance required in any embodiment of theinvention is not limited to a particular type of integrated inductor.

The details of multi-track spiral inductor design are disclosed in moredetail in U.S. patent application Ser. No. 09/493,942 filed Jan. 28,2000, (B600:36491) entitled “Multi-Track Integrated Spiral Inductor” byJames Y. C. Chang; based on U.S. Provisional Application No. 60/117,609filed Jan. 28, 1999 (B600:34072) and U.S. Provisional Application No.60/136,654 filed May 27, 1999 (B600:34676), the subject of which isincorporated in this application in its entirety by reference.

FIG. 29 is an exemplary illustration of the possible effects of inductorQ on filter selectivity in a parallel LC circuit, such as shown in 2706of FIG. 27. The Q of a spiral inductor tends to be low. In order toadvantageously control the Q so that the maximum performance of anintegrated filter may be obtained, calibration of inductor Q is used.

The overall effect of this is that when a device with high seriesresistance and thus, low Q is used as a component in a filter that theoverall filter Q is low 2902. A high Q filter response is sharper 2984.The goal of a filter is to achieve frequency selectivity. The filterselectivity is the same electrical property as selectivity in the “frontend” of the receiver previously described. If the filter has a low Qfrequencies outside the pass band of the filter will not achieve asgreat of an attenuation as if the filter contained high Q components.The high degree of selectivity is required to reject the multitude ofundesirable distortion products present in a receiver that fall close tothe tuned signal. Satisfactory inductor dimensions and device Q havebeen obstacles in integrating filters on a CMOS substrate.

Prediction of the inductance yielded by the spiral is closelyapproximated by formula. However, prediction of the inductor's Q is moredifficult. Three mechanisms contribute to loss in a monolithicallyimplemented inductor. The mechanisms are metal wire resistance,capacitive coupling to the substrate, and magnetic coupling to thesubstrate. Magnetic coupling becomes more significant in CMOStechnologies with heavily doped substrates, because the effect ofsubstrate resistance appears in parallel with the inductor. The firstfour or five turns at the center of the spiral inductor contributelittle inductance and their removal helps to increase the Q. In spite ofextensive research inductors implemented in CMOS possess Qs afterlimited to less than five.

FIG. 30 is an illustration of a typical filter bank 3002 utilized inembodiments of the invention for filtering I and Q IF signals 3208. Bandpass filters utilized in the embodiments of the invention have a centerfrequency f_(c) and are designed to provide a given selectivity outsideof the pass bond. The exemplary filters 3002 also incorporate gain. Gainand selectivity are provided by an amplification (“transconductance”)stage with an LC load, resulting in an active filter configuration thatgives the filter response shown. Selectivity is provided principally bythe LC load. The gain is attributable to the transconductance stage. Thetransconductance stage comprises a linearized differential pairamplifier that has an improved dynamic range. Over temperature thefilter response degrades as indicated in FIG. 30. This degradation istypically attributed to inductors.

With the spiral inductors utilized in the embodiments of the inventionthe gain of this filter stage is substantially determined by the Q orquality factor of the inductor. The Q is in turn substantiallydetermined by the series resistance of the metal in the spiral of theinductor. The Q decreases as temperature increases causes an increase ininductor series resistance. The decrease in Q with increasingtemperature adversely affects the filter characteristics. As can be seenin 306 at FIG. 30 as the temperature increases from 50° C. 3004 to 100°C. 3006 overall gain decreases, and selectivity is degraded.

Active Filter Utilizing a Linearized Differential Pair Amplifier

A linearized differential pair amplifier is used in the active filterspresent in the receiver. The technique utilized to linearize the CMOSdifferential pair described in light of application to active filtersmay be utilized in any application in which a differential amplifierhaving a linear response is desirable.

FIG. 31a is a diagram of an exemplary differential transconductancestage 3102 with an LC load 3104. Together the transconductance stage andLC load make up a filter 3002 that is a part of filter bank 3001. Theexemplary embodiment of the filter is disposed on a CMOS substrate thatis part of an integrated receiver.

FIG. 31b is a block diagram of a linearized differential pair amplifierthat is coupled to distortion canceling linearization circuit. Gainstage 3102 comprises a differential pair amplifier 3103 that has alinearization circuit 3105 coupled to form a linearized differentialpair. In the embodiment shown the linearization circuit is coupled inparallel to the differential pair amplifier.

The linearized differential pair typically improves maximum signalhandling capability over that of a differential pair in excess of 19 dB.In the past, typical improvements with prior art linearization schemesapplied to differential pair amplifiers tended to be around 7 dB. Thus,the approach described in the embodiment tends to have a dynamic rangeadvantage of 12 dB over the prior art.

An embodiment of the differential pair amplifier 3103 comprises a firstand second FET transistor M1 M2. Equivalently, other type of transistorare contemplated as satisfactory substitutes. A differential inputcomprises signals V_(i1) and V_(i2) coupled to the inputs of theamplifier 3103 and linearization circuit 3105. A differential outputcomprises signals V_(o1) and V_(o2).

An embodiment of the linearization circuit 3105 comprises two or moreauxiliary differential pairs 3107, 3109 respectively. Each auxiliarydifferential pair comprises a first and a second FET transistor.Auxiliary differential pair 3107 comprises transistors M3 and M4.Auxiliary differential pair 3109 comprises transistors M5 and M6.Equivalently, other type of transistor are contemplated as satisfactorysubstitutes. Further improvements in linearization is possible by addingmore auxiliary differential pairs. However, as linearization isincreased the size of transistors contained in the additional auxiliarydifferential pairs decreases. Thus, a limit in the linearization thatmay be obtained is set by the practical aspects of device matching andscaling.

FIG. 31c is an illustration depicting a representative channel of anyone of the typical field effect of transistors M1 M2, M3, M4. A channelof length l, and a width w and a thickness t is disposed on a substrateto form a field effect transistor (FET) as shown in FIG. 31c. Thechannel is provided with ohmic contacts 3111 for a drain connection anda source connection.

In an exemplary embodiment of a filter designed to operate at 275 MHzthe channel lengths of M1 M2, M3, M4, M5, and M6 were chosen to havel=0.6 μm. In Table I for an I_(ss=)9 mA and n=16 the channel widths forthe transistors in the exemplary embodiment of the 275 MHz filter areshown.

TABLE I Device Width W_(1,2) W_(4,5) W_(3,6) Iss n 1.9 um × 20 2 um × 51.95 um × 2 9 mA 16

The subscripts in table I refer to the transistor that is associatedwith a given channel width. For example W_(1,2) refers to the channelwidth of transistor M1 and M2. I_(ss) is the main pair tail currentsource, and n refers to the ratio of the main pair tail current source.

Transistor M1 and M2 has a width of 1.9 μm×20, transistor M4 and M5 havea channel width of 2.0 μ×5, and transistors M3 and M6 have a channelwidth of 1.95 μm×2. In the notation used the dimension with an “x”refers to the number of transistors coupled in parallel. For example 2.0μm×5 refers to 5 transistors with a 2 μm channel width coupled inparallel, to form an overall 10 μm channel width. An exemplary filterconstructed with these channel widths and the fixed length exhibits athird order intermodulation typically less than −70 dB when fed with atwo-tone input, each tone having a magnitude of 125 mV_(p).

The channel widths and lengths of the exemplary embodiment were chosenthrough an optimization process. The transistors in the auxiliarydifferential pair amplifiers, when stimulated by the amplifier inputwill produce a signal that when added to the gain stage output, willtend to reduce distortion.

FIG. 31d is a block diagram showing the interconnection of adifferential pair amplifier 3103 to a linearization circuit 3105. Gainstage 3103 is made up of a differential pair amplifier comprising a pairof transistors M1 and M2, each transistor having a drain, a source and agate. Transistors M1 and M2 tend to contribute to the majority of anoverall amplifier gain produced.

In the differential pair amplifier the sources of M1 and M2 are eachcoupled to a first terminal of a current source I_(ss). A secondterminal of I_(ss) is coupled to ground. Current source I_(ss) is aconventional current source implemented in a manner known to thoseskilled in the art. The drain of M1 is coupled to an output current I₁.The drain of M2 is coupled to an output comprising current I₂. Adifferential input voltage is applied across a pair of terminals V_(i1),V_(i2) that are coupled to the gates of M1 and M2, respectively.

The two auxiliary pair differential amplifiers 3107, 3109 are present asshown. The auxiliary amplifiers tend to linearize the currents I₁ andI₂. Currents 3113 and 3115 tend to subtract non-linear currents fromcurrent I₁ and current I₂ respectively. The gates of the differentialpairs 3107, 3109 are also driven by the input differential voltage thatis supplied to the differential pair amplifier 3103.

The relationship of transistor parameters of channel length and width(of FIG. 31c) in transistors M1 M2 to the transistor parameters of M3,M4, M5, M6, contained in the auxiliary differential pair amplifiers 3107and 3109 of the linearization circuit, is to minimize distortion. Thetransistors function in relation to each other such that distortioncreated by the transistors in the differential pair amplifier generatingcurrent outputs I₁ and I₂ tends to be reduced by the currents generatedby the transistors M3, M4, M5, M6 of the auxiliary differential pairamplifiers 3113, 3115. In order to select appropriate transistorparameters a new CMOS differential pair linearization technique isutilized. The technique is found from examining the operating parametersof a differential pair amplifiers and cross coupled differential pairamplifiers.

FIG. 31e is a schematic illustrating a CMOS differential pair oftransistors. In the exemplary embodiment the transistors are biased tooperate in the saturation region. The differential pair of transistorsgenerate a differential current output I_(d1) and I_(d2) that isproportional to a differential input voltage, supplied by a pair ofvoltages V_(i1) and V_(i2) as referenced to a circuit ground potential.The differential pair of transistors is comprised of a first transistorM1 and a second transistor M2.

Each transistor M1, M2 has a drain, a source and a gate terminal. Thesources of M1 and M2 are coupled to a first terminal of a current sourceI_(ss). The current source I_(ss) has a second terminal which is coupledto the circuit ground. Current source I_(ss) is constructedconventionally as is known to those skilled in the art. The voltagesV_(i1) and V_(i2) are applied to the gates of transistors M1 and M2respectively. The drains of transistors M1 and M2 supply the currentoutputs I_(d1) and I_(d2) respectively.

The differential pair of FIG. 31e is biased so that each transistor M1and M2 operates in the saturation region defined by(V_(GS)−V_(th))_(M1,2)≦V_(DS) for each transistor M1 and M2 Derivationof this relationship is disclosed in “Analysis and Design of AnalogIntegrated Circuit Design”, by P. R. Gray and R. G. Meyer, 3^(rd) ed.John Wiley and Sons, 1983, the disclosure of which is hereinincorporated in its entirety by reference. Where V_(GS) is a gate sourcevoltage as measured across the gate and source terminals of M1 and M2,V_(DS) is a drain source voltage as measured across the drain and sourceterminals of M1 and M2, and V_(th) is a threshold voltage associatedwith M1 and M2. A derived term V_(gt) is defined in conjunction withequation (7.1) and is equal to on V_(gs)−V_(th). The superscriptnotation M1,2 associated with V_(gt) indicates the parameter isassociated with transistors M1 and M2. When the differential pair shownin FIG. 31e is biased in the saturation region the current and voltagerelationship is given by equation (7.1). $\begin{matrix}{{\Delta \quad I_{d}} = {{I_{ss} \times \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} \times \{ {1 - {\frac{1}{4}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{2}}} \}^{0.5}\quad \text{for}\quad \Delta \quad V_{i}} \leq {\sqrt{2} \times V_{gt}^{{M1},2}}}} & \text{(7.1)}\end{matrix}$

where:

ΔI_(d)=I_(d1)−I_(d2)

ΔV_(i)=V_(i1)−V_(i2)

V_(gt) ^(M1,2)=(V_(GS)−V_(th))_(M1,2)@ΔV_(i)=0

Note that ΔV_(i) denotes the peak signal level for each of the twosignals.

A series expansion for (1−x²)^(0.5) is applied to equation (7.1) toobtain equation (7.2) as a current output defined in terms of a sum of aseries of input voltages each raised to progressively greaterexponential powers. $\begin{matrix}{{\Delta \quad I_{d}} = {I_{ss} \times \{ {( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} ) - {\frac{1}{8}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{3}} - {\frac{1}{128}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{5}} - {\frac{1}{1024}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{7}\quad \ldots}}\quad \}}} & \text{(7.2)}\end{matrix}$

For small input signals ΔV_(i) satisfying the condition, ΔV_(i)<<ΔV_(gt)^(M1,2), the first linear term of equation 2 is much larger compared tothe higher order terms. Under this condition, the output current ΔI_(d)is almost a linear function of input voltage ΔV_(i).

However, as the input signal level approaches V_(gt) ^(M1,2) higherorder terms tend to contribute more to the output current. Thecontribution of the higher order, nonlinear terms gives rise to spuriousharmonic components and intermodulation distortion (IM3). Thus thedifferential amplifier behaves linearly for small input signals andbegins to distort when large signals are applied.

In filter design the more significant spurious response tends to bethird order intermodulation distortion. The following process forminimizing distortion is carried out by considering only intermodulationdistortion present in a differential pair amplifier.

For the differential pair of FIG. 31e, third order intermodulationdistortion (IM3) is given in equation (7.6).

To calculate IM3, the coefficients in the following equation must firstbe found:

ΔI _(d) =a ₁ v _(i) +a ₂ v _(i) ² +a ₃ v _(i) ³ +a ₄ v _(i) ⁴ +a ₅ v_(i) ⁵ +a ₆ v _(i) ⁶+ . . .   (7.3)

Where v_(i) denotes the input voltage.

By comparing equation (7.3) to equation (7.6) the coefficients ofequation (7.3) are determined: $\begin{matrix}\begin{matrix}{{a_{1} = \frac{I_{ss}}{( {V_{GS} - V_{th}} )}}\quad} & {{a_{2} = 0}} \\{a_{3} = {- \frac{I_{ss}}{8( {V_{GS} - V_{th}} )^{3}}}} & {{a_{4} = 0}} \\{a_{5} = {- \frac{I_{ss}}{128( {V_{GS} - V_{th}} )^{5}}}} & {{a_{6} = 0}}\end{matrix} & \text{(7.4)}\end{matrix}$

The third order intermodulation components IM3, are known to begenerated by the odd coefficients Thus, by collecting the terms havingodd coefficients, and defining their sum to be the third orderintermodulation (“IM3”) the following equation (7.5) is obtained.

Peak input voltage is denoted by a caret over the letter {circumflexover (v)}_(i). $\begin{matrix}{{IM3} \approx {{\frac{3}{4}\frac{a_{3}}{a_{1}}{\hat{v}}_{i}^{2}} + {\frac{25}{8}\frac{a_{5}}{a_{1}}{\hat{v}}_{i}^{4}} + \ldots}} & \text{(7.5)}\end{matrix}$

Inserting the values for a₁ and a₃ and a₅ from eq 4.22 into equation(4.23) yields an expression for third order intermodulation (IM3) thatis expressed in terms of a differential pair amplifiers transistorparameters. $\begin{matrix}{{IM}_{3} \approx {{\frac{3}{32} \times ( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{2}} + {\frac{25}{1024}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{4}} + {\frac{735}{2^{16}}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{6}} + \ldots}} & \text{(7.6)}\end{matrix}$

The ΔV_(i) of FIG. 31e denotes peak signal level for each of the twoinput signals.

A large signal transconductance (“G_(m)”) is the rate of change of inputcurrent (ΔI_(d)) with respect to the rate of change of the input voltage(ΔV_(i)). Large signal transconductance is found by differentiatingequation (7.2) with respect to ΔV_(i) to yield an expression for largesignal transconductance. $\begin{matrix}\begin{matrix}{G_{m} = \frac{{\Delta}\quad I_{d}}{{\Delta}\quad V_{i}}} \\{\approx {\frac{I_{ss}}{V_{gt}^{{M1},2}}\{ {1 - {\frac{3}{8} \times ( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{2}} - {\frac{5}{128}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{4}} -} }} \\ {\frac{7}{1024}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{6}\quad \ldots} \}\end{matrix} & \text{(7.7)}\end{matrix}$

The first term of equation (7.7) represents a small signaltransconductance (“g_(m)”): $\begin{matrix}{g_{m} = \frac{I_{ss}}{V_{gt}^{{M1},2}}} & \text{(7.8)}\end{matrix}$

A deviation of large signal transconductance (G_(m)) from small signaltransconductance (g_(m)) is defined to be: $\begin{matrix}{\frac{\Delta \quad G_{m}}{g_{m}} \approx {{{- \frac{3}{8}} \times ( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{2}} - {\frac{5}{128}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{4}} - {\frac{7}{1024}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{6}\quad \ldots}}} & \text{(7.9)}\end{matrix}$

Transconductance variations are given by equation (7.9) which representsa fractional change in transconductance for variations in input signallevel.

By examining the equations derived for relatively small signals, arelationship between two of the equations is noted. There is arelationship between the equation for third order intermodulationdistortion and the equation for transconductance variations. Theequations have common terms, and are directly proportional to eachother. For a given input level, on examining equations and the thirdorder intermodulation distortion level of equation (7.6) isapproximately one-quarter of the transconductance variations of equation(7.7). This relationship for small signals is expressed in equation(7.10): $\begin{matrix}{{IM}_{3} \approx {\frac{1}{4} \times \frac{\Delta \quad G_{m}}{g_{m}}\quad \text{for}\quad \Delta \quad V_{i}\quad {\langle{\langle\quad V_{gt}^{{M1},2}}}}} & \text{(7.10)}\end{matrix}$

The relationship in equation (7.10) suggests that third orderintermodulation distortion is controlled by controlling variations intransconductance that typically occur when the input voltage changes.Thus, to a first order of approximation, minimization oftransconductance variations over a range of input signal levels tends toreduce third order intermodulation distortion (IM3). The minimization oftransconductance variations is found by applying calculus to minimizethe functional relationship.

FIG. 31f is a graph of a differential current (ΔI_(1,2)=ΔI_(d)) andnormalized transconductance (G_(m)/g_(m)) as input voltage(V_(in)=ΔV_(i)) is varied in the differential pair of FIG. 31e. Fromthis curve an exemplary baseline intermodulation distortion for anuncompensated differential pair amplifier of FIG. 31e is found. Increating this graph values of, V_(gt) ^(M1,2)=0.7V and I_(ss)=2.4 mAwere used. The graph shows the increasing non-linearities present in theoutput current (ΔI_(1,2)=ΔI_(d))as the input voltage (V_(in)=ΔV_(i))driving the amplifier increases.

For an input voltage of 250 mv the large signal transconductance is 0.96times the small signal transconductance 3117. Thus, ΔG_(m)/g_(m)≈0.04.By substituting 0.04 into equation (7.10) the third order IM level is{fraction (1/100)}, or −40 dB (−40=20 Log ({fraction (1/100)})). Adifferential pair amplifier comprises a baseline from which improvementsin linearity are measured. Interconnected linearizing circuitry is nextadded to the differential pair amplifier of FIG. 31e to improve itslinearity.

FIG. 31g is a schematic diagram of a differential pair amplifier 3127with a second cross coupled differential pair error amplifier 3129 addedthat tends to reduce distortion.

Linearity of a differential pair amplifier may be improved by usinglarge values of an applied gate overdrive voltage (V_(GS)−V_(th))_(M1,2)that is applied to transistors M1 and M2. A limiting factor in utilizinglarge values of gate overdrive voltage is a maximum available supplyvoltage. With a reduced scaling of device sizes common in today's morecompact circuit layouts, a maximum available supply of voltage tends tobe reduced. Since a higher voltage required for a gate overdrivecondition is not present, alternative linearization techniques aredesirable. One technique is the addition of a cross-coupled differentialpair 3129, that functions as an error amplifier, to a differential pairamplifier 3127.

A preferable linearization process takes the form of adding errorcurrents I_(d3) I_(d4) to differential amplifier currents Id₁ I_(d2) ina way that tends to improve the linearity of output currents 3131 3133.The error currents I_(d3) and I_(d4) are subtracted tend to becomenon-linear more rapidly than the currents of the differential pairamplifier I_(d1) and I_(d2).

Subtraction is achieved by cross coupling the amplifiers 3127 and 3129.A differential signal may be referenced to ground by considering it tobe made up of two signals. The equivalent signal is a set of twoindividual signals, 180 degrees out of phase and of equal amplitudereferenced to ground. In a differential voltage signal the voltages haveopposite polarities of equal amplitude at any given time.

In a differential current signal the currents flow in oppositedirections and are of equal magnitude at any given time. In the case ofa current one signal flows into the terminal, the other out of it. Ifthe two differential signals are coupled to the same terminal theresultant signal would be canceled since each signal is equal andopposite. If the signals are unequal the cancellation is not total.

Thus, by cross coupling the differential pair amplifier 3127 to theerror amplifiers 3129 in parallel the currents I_(d3) I_(d4) present ineach drain of the error amplifier are coupled to the drain currentsI_(d2) I_(d1) of the differential pair amplifier respectively. Pairedsignals I_(d3) I_(d2) and I_(d4) I_(d1) are 180 degrees out of phase andunequal in amplitude, causing a subtraction of The error amplifiercurrent from the differential pair amplifier current in each lead.

The differential pair amplifier 3127 has a differential inputV_(i1 and V) _(i2). The differential pair amplifier has a differentialcurrent output provided by currents 3131 and 3133. By Kirchhoff'scurrent law the current 3133 flowing out of node 3121 is equal to a sumof branch currents I_(d3) and I_(d2) into node 3121. Similarly, current3131 flowing out of node 3119 is equal to a sum of branch currentsI_(d1) and I_(d4) flowing into node 3119. To provide the branch currentsa main differential pair 3127 and an auxiliary differential amplifieralternatively termed an error amplifier 3129 are provided.

The main differential pair 3127 comprises transistors M1 and M2. Thegates of transistors M1 and M2 are driven by differential input voltageV_(i1) and V_(i2). The sources of M1 and M2 are coupled to a firstterminal of a conventional current source I_(ss). A second terminal ofI_(ss) is coupled to ground. The drains of M1 and M2 provide outputcurrents I_(d1) and I_(d2), respectively.

The auxiliary cross-coupled differential pair 3129 comprises transistorsM3 and M4. The gate of M3 is coupled to the gate of M1 and the gate ofM4 is coupled to the gate of M2. The sources of M3 and M4 are coupledtogether. The coupled sources of M3 and M4 are in turn coupled to afirst terminal of a current source I_(ss)/n. Current from sourceI_(ss)/n is a fraction of I_(ss) in order to control the current outputI_(d3) I_(d4) of The auxiliary amplifier. A second terminal of I_(ss)/nis coupled to ground. The drain of M3 is coupled to the drain of M2. Thedrain of M4 is coupled to the drain of M1. This connection of gates anddrains creates the desired cross coupling.

The current and voltage relationships in the cross coupled differentialamplifier are as follows:

where:

ΔI_(d) ^(1,2) =ΔI _(d) ^(3,4) =ΔI _(Total) =ΔI _(d) ^(1,2) −VI _(d)^(3,4)  (7.11)

The ΔI_(d) ^(1,2) is given by: $\begin{matrix}\begin{matrix}{{\Delta \quad I_{d}^{1,2}} = {{\Delta \quad I_{d}} = {I_{d1} - I_{d2}}}} \\{= {I_{ss} \times \{ {( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} ) - {\frac{1}{8}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{3}} - {\frac{1}{128}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{5}} -} }} \\{ {\frac{1}{1024}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{7}\quad \ldots}\quad \}}\end{matrix} & \text{(7.12)}\end{matrix}$

The ΔI_(d) ^(3,4) is given by: $\begin{matrix}\begin{matrix}{{\Delta \quad I_{d}^{3,4}} = {I_{d3} - I_{d4}}} \\{= {\frac{I_{ss}}{n} \times \{ {( \frac{\Delta \quad V_{i}}{V_{gt}^{{M3},4}} ) - {\frac{1}{8}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M3},4}} )^{3}} - {\frac{1}{128}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M3},4}} )^{5}} -} }} \\{ {\frac{1}{1024}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M3},4}} )^{7}\quad \ldots}\quad \}}\end{matrix} & \text{(7.13)}\end{matrix}$

Assuming that $\frac{V_{gt}^{{M1},2}}{V_{gt}^{{M3},4}} = m$

and thus ${\frac{W^{{M1},2}}{W^{{M3},4}} = \frac{n}{m^{2}}},$

the total current is found to be: $\begin{matrix}{{\Delta \quad I_{Total}} = {I_{ss} \times \{ {{( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )( {1 - \frac{m}{n}} )} - {\frac{1}{8}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{3}( {1 - \frac{m^{3}}{n}} )} - {\frac{1}{128}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{5}( {1 - \frac{m^{5}}{n}} )} - {\frac{1}{1024}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{7}( {1 - \frac{m^{7}}{n}} )\quad \ldots}}\quad \}}} & \text{(7.14)}\end{matrix}$

Where the ration of the channel widths comes from the current sourceshaving a ratio of n, and the v_(gt)s have a ratio of m. Thus, for a MOStransistor operating in saturation: $\begin{matrix}{I_{ds} = {\frac{\mu \quad {Cox}\quad W}{2L}( {V_{gs} - V_{th}} )^{2}}} & \text{(7.15)} \\{\frac{I_{ds}^{{M1},2}}{I_{ds}^{{M3},4}} = \frac{{W^{{M1},2}( ( {V_{gs} - V_{th}} )^{2} )}^{{M1},2}}{{W^{{M3},4}( ( {V_{gs} - V_{th}} )^{2} )}^{{M3},4}}} & \text{(7.16)} \\{{->\frac{W^{{M1},2}}{W^{{M3},4}}} = \frac{n}{m^{2}}} & \text{(7.17)}\end{matrix}$

The third order term of equation (7.14) that controls the contributionof third order intermodulation goes to zero when m³/n=1. The crosscoupled differential amplifier is described in more detail in P. R. Grayand R. G. Myer, “Analysis and Designs of Analog Integrated CircuitDesign,” Third Edition, John Wiley & Sons, 1993. Utilizing a value ofn=9.5 and m=2, a dynamic range of the input to the amplifier isincreased by 6.5 dB, for an IM3 level of −40 dB. Where n is the ratio ofcurrent source values, and the ratio of m to n was previously defined.

The dynamic range of the input to maintain a −40 dB third orderintermodulation level may be further extended. Extension of dynamicrange is possible by using two or more differential pairs cross-coupledin parallel to a main differential pair. In an embodiment, the maindifferential pair has two auxiliary differential pairs associated withit to linearize the main differential pairs output.

FIG. 31h is a graph illustrating The linearized output current of across coupled differential output amplifier. The auxiliary differentialpair amplifier 3129 of FIG. 31g subtracts a small current I_(d3), I_(d4)from the output current of the differential pair amplifier ΔI_(d)^(3,4). The currents I_(d3), I_(d4) are subtracted from The outputcurrents I_(d1) and I_(d2), respectively. This small amount of currenttends to become nonlinear more rapidly than ΔI_(d) ^(1,2).

The derivation above for the circuit of 31 h utilized a ratio of channelwidths to adjust The proper error amplifier currents to cancel the thirdorder intermodulation distortion. A chosen channel width for transistorsM1 and M2 was selected, and a channel width was found for transistors M3and M4 that tends to yield an IM3 level of −40 dB. This yields anincrease in dynamic range of approximately 6.5 dB. Increasing the numberof auxiliary differential pairs present and utilizing a linearizationoptimization process tends to improve overall amplifier linearity.

FIG. 31i is a schematic of a differential pair amplifier 3102incorporating two auxiliary cross-coupled differential pairs 3107 3109to improve linearization of the output response I₁ and I₂. The maindifferential pair 3103 comprises transistors M1 and M2. The gates of M1and M2 are coupled to a differential input voltage V_(i1 and V) _(i2).The sources of M1 and M2 are coupled to a first terminal of currentsource I_(ss). A second terminal of I_(ss) is coupled to ground. Currentsource I_(ss) is typically constructed as known to those skilled in theart. The drains of M1 and M2 supply currents I_(d1) and I_(d2),respectively. The drains of M1 and M2 are coupled to current outputs I₁and I₂, respectively.

The first auxiliary differential pair 3107 comprises transistors M3 andM4. The gate of M3 is coupled to differential input voltage V_(i1). Thegate of M4 is coupled to differential input voltage V_(i2). The sourcesof M3 and M4 are coupled together and then to a first terminal of afirst current source I_(ss)/n. A second terminal of I_(ss)/n is coupledto a ground potential. Current source I_(ss)/n is typically constructedas a conventional current source as is known to those skilled in theart. The drain of M3 is coupled to the drain of M2. The drain of M4 iscoupled to the drain of M1.

The second auxiliary differential pair 3109 comprises transistors M5 andM6. The gate of M5 is coupled to differential input voltage V_(i1). Thegate of M6 is coupled to differential input voltage V_(i2). The sourcesof M5 and M6 are tied together to a first terminal of a second currentsource I_(ss)/n. A second terminal of I_(ss)/n is coupled to ground. Thesource of M5 is coupled to the source of M2. The source of M6 is coupledto the source of M1.

FIG. 31j is a graph of the currents present in the main and twoauxiliary differential pair amplifiers graphed against input voltage asmeasured across the input terminals where Vin=V_(i1)−V_(i2). This graphillustrates an offset between currents ΔI_(3,4) and ΔI_(5,6). An offsetis present where the input voltage passes through zero 3135. Thecurrents ΔI_(3,4) and ΔI_(5,6) from the auxiliary differential pairamplifiers are much smaller than the main differential pair amplifiercurrent ΔI_(1,2). It is desired to produce an output current that variesa linear relationship to the input voltage. The differential currentsfrom the auxiliary differential pairs ΔI_(5,6) and I Δ_(3,4) aresubtracted from ΔI_(1,2) to produce curve of total differential outputcurrent ΔI_(Total).

The composite curve ΔI_(Total) is a more linear curve than ΔI_(1,2).Thus, by subtracting the currents produced by the auxiliary differentialpair amplifiers, The linearity of The current versus voltage response isimproved. The amount of current produced in auxiliary cross-coupleddifferential pairs over a range of input voltage Vin is related to atransconductance characteristic of each of a set of transistors in theamplifier.

Thus, to shape The ΔI_(Total) curve is necessary to fabricate M3, M4,M5, and M6 so that the currents they produce will contribute to Thelinearization of The ΔI_(Total) curve. Shaping is done throughmanipulating transconductance. Transconductance is an inherenttransistor parameter related to drain current I_(d). It is defined asfollows: $\begin{matrix}{g_{m} = \frac{I_{d}}{V_{gs}}} & \text{(7.18)}\end{matrix}$

Thus, by controlling the transconductance of the transistors in theauxiliary differential pairs, the output current of the maindifferential pair is linearized by superposition of the currents. Toreduce third order inter modulation close to zero, a flat G_(m) curvefor the amplifier tends to be advantageous.

FIG. 31k is a graph of transconductance curves for the differentialamplifier made up of a main differential pair amplifier 3103 and alinearization circuit 3105 comprising differential pair amplifiers 3107and 3109. The main differential pair amplifier possesses atransconductance characteristic shown by the curve G_(m) ^(M1,2) havinga peaked response.

To reduce third order air modulation distortion, it is desirable toshape the transconductance curve G_(m) ^(M1,2) so that the peak of isflattened as shown by the curve G_(m) ^(Total). flattening isaccomplished by subtracting or decreasing the G_(m) in the peak regionof the curve. The decrease is achieved by The linearization circuit3105.

Auxiliary differential pair amplifier 1 3107 exhibits a characteristictransconductance curve centered about a voltage offset V_(os) from zeroinput volts, and is denoted G_(m) ^(M3,4) on the graph. Thetransconductance curve for auxiliary differential pair amplifier 2 3109is offset in the negative direction from zero input voltage by an amountthat is equal to the first auxiliary pair V_(os), this curve is denotedG_(m) ^(M5,6).

FIG. 311 illustrates an equivalent circuit that provides an offsetvoltage V_(os) that permits shaping of The G_(m) ^(Total) curve. Theaddition of an offset voltage in The auxiliary differential pairamplifiers allows a more accurate cancellation of non-linearities. Theintroduction of offset voltage V_(os) is illustrated by the addition ofa voltage source in the gate leads of M3 and M6. The voltage source addsin series with The input voltages V_(i1) and V_(i2) to create theoffset. The voltage source is shown as a battery. However, the offsetvoltage is equivalently added in a number of ways comprising building itinto the semiconductor circuit parameters and providing biasingcircuitry. The offset voltages are built into the circuit by choosingdifferent widths for the auxiliary differential pair devices.

Returning to FIG. 31k, the transconductance curves of the auxiliarydifferential pairs add to form a G_(m) curve shown by G_(m)^(M3,4)+G_(m) ^(M5,6). The composite curve G_(m) ^(M3,4)+G_(m) ^(M5,6)is subtracted from the main differential pair curve G_(m) ^(M1,2) toproduce a final composite transconductance curve G_(m) ^(Total) thatcontrols the overall amplifier current response and linearity. Thecurrent relationships for a differential pair amplifier that includesoffsets in The linearization circuit are as follows:

G _(m) ^(Total) =G _(m) ^(M1,2)−(G _(m) ^(M3,4) +G _(m) ^(M5,6))  (7.19)

The current in the auxiliary pairs is given by: $\begin{matrix}\begin{matrix}{{\Delta \quad I_{d}^{3,4}} = {I_{d3} - I_{d4}}} \\{= {\frac{I_{ss}}{n} \times \begin{Bmatrix}{( \frac{{\Delta \quad V_{i}} + V_{os}}{V_{gt}^{{M3},4}} ) - {\frac{1}{8}( \frac{{\Delta \quad V_{i}} + V_{os}}{V_{gt}^{{M3},4}} )^{3}} -} \\{{{\frac{1}{128}( \frac{{\Delta \quad V_{i}} + V_{os}}{V_{gt}^{{M3},4}} )^{5}} - {\frac{1}{1024}( \frac{{\Delta \quad V_{i}} + V_{os}}{V_{gt}^{{M3},4}} )^{7}\quad \ldots}}\quad}\end{Bmatrix}}}\end{matrix} & \text{(7.20)} \\\begin{matrix}{{\Delta \quad I_{d}^{5,6}} = {I_{d5} - I_{d6}}} \\{= {\frac{I_{ss}}{n} \times \begin{Bmatrix}{( \frac{{\Delta \quad V_{i}} - V_{os}}{V_{gt}^{{M5},6}} ) - {\frac{1}{8}\quad ( \frac{{\Delta \quad V_{i}} - V_{os}}{V_{gt}^{{M5},6}} )^{3}} -} \\{{{\frac{1}{128}( \frac{{\Delta \quad V_{i}} - V_{os}}{V_{gt}^{{M5},6}} )^{5}} - {\frac{1}{1024}( \frac{{\Delta \quad V_{i}} - V_{os}}{V_{gt}^{{M5},6}} )^{7}\quad \ldots}}\quad}\end{Bmatrix}}}\end{matrix} & \text{(7.21)}\end{matrix}$

And the total current is: $\begin{matrix}\begin{matrix}{{\Delta \quad I_{Total}} = {{\Delta \quad I_{d}^{1,2}} - ( {{\Delta \quad I_{d}^{3,4}} - {\Delta \quad I_{d}^{5,6}}} )}} \\{= {{I_{ss} \times \{ {( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} ) - {\frac{1}{8}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{3}} - {\frac{1}{128}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{5}} - {\frac{1}{1024}( \frac{\Delta \quad V_{i}}{V_{gt}^{{M1},2}} )^{7}\ldots}} \}} -}} \\{{{\frac{I_{ss}}{n} \times \{ {( \frac{{\Delta \quad V_{i}} - V_{os}}{V_{gt}^{{M5},6}} ) - {\frac{1}{8}( \frac{{\Delta \quad V_{i}} - V_{os}}{V_{gt}^{{M5},6}} )^{3}} - {\frac{1}{128}( \frac{{\Delta \quad V_{i}} - V_{os}}{V_{gt}^{{M5},6}} )^{5}} - {\frac{1}{1024}( \frac{{\Delta \quad V_{i}} - V_{os}}{V_{gt}^{{M5},6}} )^{7}\ldots}} \}} -}} \\{{\frac{I_{ss}}{n} \times \{ {( \frac{{\Delta \quad V_{i}} + V_{os}}{V_{gt}^{{M3},4}} ) - {\frac{1}{8}( \frac{{\Delta \quad V_{i}} + V_{os}}{V_{gt}^{{M3},4}} )^{3}} - {\frac{1}{128}( \frac{{\Delta \quad V_{i}} + V_{os}}{V_{gt}^{{M3},4}} )^{5}} - {\frac{1}{1024}( \frac{{\Delta \quad V_{i}} + V_{os}}{V_{gt}^{{M3},4}} )^{7}\ldots}} \}}}\end{matrix} & (7.22)\end{matrix}$

The desired end result is to choose variables Vos, V_(gt) ^(M4,4,5,6)and n for equation (7.22) so that a plot of ΔI_(Total) verses V_(in)results in a straight line. An optimization package to aid calculationsis equivalently utilized to determine the desired parameters. A straightline has constant slope. The slope of the line is found by taking thefirst derivative. For The best possible linearity equation (7.22) isdifferentiated with respect to input voltage. Equation (7.22) issymmetrical with respect to input voltage. Thus, the even orderderivative terms are set to zero when evaluated at zero input voltage.Next, optimal values are derived for the three parameters n, V_(os), andV_(gt) ^(M3,4,5,6). The result is a maximally flat transconductancecurve that yields a linear current verses voltage curve.

For example in a design that requires an IM3 better than 65 dB isrequired. From equation (7.10) a transconductance curve to achieve thedesired IM3 has a flatness tending to be no greater than +/−0.25 dB. Tofind the desired values the optimization process is carried out byinspection coupled with a process of trial and error. In using aniterative optimization process the following values were selected as astarting point:

V _(gt) ^(M3,4) ≈V _(gt) ^(M1,2)/2 V _(OS) ≈V _(gt) ^(M3,4)/3  (7.23)

The offset voltages are built into the integrated circuit by choosingthe W/L ratio so that transistors that comprise the same differentialpair have differing widths. For example as previously shown in Table I.In the case of a linearization circuit 3105, as shown in FIG. 311 theW/L ratio of M3 and M6 is different from M4 and M5. $\begin{matrix}{V_{os} \approx {( \frac{V_{gt}^{{M3},4}}{2} ) \times \frac{\Delta ( {W/L} )}{W/L}}} & (7.24)\end{matrix}$

The widths are found from equations (7.23) and (7.24). This completes afirst pass of The design. Next the simulation program is utilized. Inthe simulation transconductance verses voltage and transistor channelwidths are optimized to yield the targeted flatness.

In an alternative embodiment, a high degree of linearity is not benecessary. Ripple is allowed in the transconductance curve to producesatisfactory linearity.

In an embodiment the maximally flat transconductance curve for smallsignals zero IM3 distortion is produced. However, if the curve must bemaximally flat, the range of values for V_(in) is reduced. In thealternative embodiment allowing some ripple in the transconductancecurve, allows the range of input voltage V_(in) is that produces afinite intermodulation distortion to be extended.

FIG. 31m is a graph of the transconductance curve for The exemplarydifferential pair amplifier that extends the input voltage range byallowing ripple in the overall G_(m) of the amplifier. Thetransconductance curve for a single differential pair amplifier 3137 iscompared to one of FIG. 31i that utilizes The parameters of Table I3139. By allowing ripple in the transconductance the range of V_(in) hasbeen extended.

In an embodiment a number of additional auxiliary differential pairs areadded to control IM3 distortion. However, if the devices required toimplement the function obtained for a given linearity are too small thanthe amplifier cannot be built successfully.

Table II compares to tone intermodulation distortions simulation resultsfor a differential pair against a structure described in an embodimentof the invention.

TABLE II Two Tone Intermodulation Distortion Simulation Results Vi_peakeach 40 mV 50 mV 100 mV 200 mV 250 mV 300 mV 350 mV Simple diff. −73 dB−69.5 dB −57 dB −45 dB −41 dB −37 dB −34.5 dB pair New structure −80 dB  −80 dB −75 dB −73 dB −73 dB −73 dB   −57 dB

Initially, at a 40 mV peak input strength for each of two signals inputto the amplifier, linearity in the embodiment is improved to −80 dB. Atapproximately, a 100 mV input signal strength, the difference in intermodulation between the prior art structure and the embodiment of theinvention approaches 20 dB. The amplifier provides a layer response upto approximately a 350 mV peak input signal. Extending the linear inputrange by approximately 12 dB results in four times the signal handlingcapability of that available in the prior art.

The details of linearizing a CMOS differential pair are disclosed inmore detail in U.S. patent application Ser. No. 09/573,356, filed May17, 2000, (B600:36523) entitled “System and Method for Linearizing aCMOS Differential Pair” by Haideh Khorramabadi; based on U.S.Provisional Application No. 60/136,115 filed May 26, 1999 (B600:34678),the subject of which is incorporated in this application in its entiretyby reference.

FIG. 32 shows a transconductance stage 3102 with an LC load 3104 that isprovided with Q enhancement 3202 and Q compensation over temperature3206. Q enhancement 3202 tends to increase the circuit Q thus,increasing the frequency selectivity of the circuit. A Q enhancement isprovided by the transconductance element's G_(m), 3202 connected asshown. Addition of this transconductance element is equivalent to addinga negative resistance 3024 that is temperature dependent in parallelwith R′(T). This negative resistance tends to cause cancellation of theparasitic resistance thus, tending to increase the circuit Q.

The details of Q enhanced filters are disclosed in more detail in U.S.patent application Ser. No. 09/573,356, filed May 17, 2000(B600: 36523)entitled, “System and Method for Linearizing a CMOS Differential Pair”by Haideh Khorramabadi; based on U.S. Provisional Application No.60/136,115 filed May 26, 1999 (B600:34678), the subject matter of whichis incorporated in this application in its entirety by reference. Oncean improved Q is achieved it is desirable to maintain it over the rangeof temperatures encountered in a circuit operation with temperaturecompensation circuitry 3206.

Due to a large positive temperature coefficient inductor quality factor(Q) is proportional to temperature. As temperature increases theresistance in the spiral increases, degrading the Q. The addition oftransconductance from the G_(m) stage 3102 tends to increase the Q ofthe filter. However, the effects of temperature on quality factor tendsto cause wide gain variation tending to need further improvement. In anembodiment of the invention for a temperature range from 0 to 100° C., Qand gain vary +/−15% in an unenhanced filter. In an embodiment with a Qenhanced filter, the Q and gain variation is doubled. In multiple stagesof filtering used in the embodiments, over 20 db of gain variation isthus encountered over temperature with the Q enhanced filters. Thisresults in an unacceptable change in the conversion gain of thereceiver. A further means of reducing the variation in Q (and thus gain)over temperature is desirable 3206.

Active Filter Inductor Q Temperature Compensation

FIG. 33 shows a method of stabilizing inductor Q over temperature 3206.This method advantageously uses a DC calibration loop 3202 and a dummyinductor 3304 to control the value of inductor series resistance R(T)and a resistive element R(1/T) 3314 to produce a net constantresistance. Thus, Q induced variation in filter response due totemperature are controlled. This method advantageously does not requirethe use of any high frequency signals in the tuning process. An inductor3306 as utilized in the filters of FIG. 30's filter bank 3002 with itsassociated series resistance R(T) is shown as an element in atemperature compensation circuit 3208. An electronic device thatsupplies a variable resistance 3310 of an amount inversely proportionalto temperature is added into the circuit 3314. The decreasing resistanceof the additional resistance 3314 with increasing temperaturecounteracts the increasing resistance of the inductor's seriesresistance R(T). In the circuit diagram this decreasing resistance isshown schematically as R(1/T). This resistance is provided by the activeresistance of a PMOS transistor biased accordingly 3314. However anydevice capable of producing the desired resistance characteristicdescribed above is an acceptable substitute.

A PMOS resistor is used in two places 3312,3314 to place the controlelement 3314 in the circuit and remove the control circuit 3208 from amain circuit 3308. In the embodiment shown, the PMOS transistor's gateto source connection is placed in series with the spiral inductor 3306of the LC circuit 3308 making up an active filter stage. The activefilter stage is controlled from a remotely located control circuit 3208that contains a duplicate PMOS resistor 3312 and inductor 3304. Inductor3304 is advantageously fabricated with the same mask pattern as used forinductor 3306. The control circuitry 3208 is not a part of the filtercircuitry 3308 in order to prevent undesirable interactions with theradio frequency signals present in the filter. In the control circuitshown, the active resistor 3312 in series with the spiral inductor 3304is duplicated remotely from the filter circuit 3308. To communicate thecontrol signal 3316 the gate of the PMOS resistor 3312 is coupled to thegate of the PMOS resistor in the filter 3314.

The control circuit provides a conventional constant current and aconventional constant voltage source function to maintain a constantcurrent through and voltage across the dummy spiral inductor 3304duplicated in the control circuit. An exemplary constant current andconstant voltage source is shown 3302 incorporating a dummy inductor3304. However, any circuit that maintains a constant voltage across, andcurrent through the inductor 3304 in the control circuit 3208 issufficient for the design.

As gate voltage 3316 changes to maintain the constant current andvoltage across the inductor in the control circuit 3304, the gatecontrol signal 3316 is simultaneously fed to the LC filter stage 3308PMOS transmitter 3314 to control the resistance, and thus the Q, of theinductor in the filter circuit 3308.

An exemplary constant current and voltage source is illustrated 3302comprising dummy inductor 3304. A temperature independent voltagereference V_(ref) is established by resistor R and conventional currentsources I. Amplifier A's negative input is connected to the voltagereference, and its positive input is connected to a symmetrical pointbetween an identical current source and the dummy inductor. The outputof amplifier A is fed into the gate of the transistor functioning as avariable resistor 3312. The constant voltage drop over temperature atthe node V_(ref) is compared to the voltage at the positive amplifierterminal. The amplifier controls the resistance of the PMOS transistorso that a constant current and constant voltage are maintained acrossthe dummy inductor.

The calibration of inductor Q is described in more detail in U.S. patentapplication Ser. No. 09/439,156 filed Nov. 12, 1999 (B600:34014)entitled “Temperature Compensation for Internal Inductor Resistance” byPieter Vorenkamp, Klaas Bult and Frank Carr; based on U.S. ProvisionalApplication No. 60/108,459 filed Nov. 12, 1998 (B600:33586), the subjectmatter of which is incorporated in its entirety by reference.

Communications Receiver

FIG. 34 is a block diagram of a communications network utilizing areceiver 3402 according to an exemplary embodiment of the invention. Acommunications network, such as a cable TV network 3404, capable ofgenerating signals provides radio frequency (“RF”) signals 3406 over theair waves, through a cable or other transmission medium. Such a signalis typically single ended, although differential transmission iscontemplated. A receiver front end 3408 next converts the RF singleended signal to a differential signal. In the embodiment shown the frontend provides low noise amplification of a weak received signal by a lownoise amplifier. The embodiment shown also includes an attenuator toreduce a strong received signal's level. An externally supplied controlsignal 4302 controls the amount of attenuation, or gain of the RFsignal. A receiver front end, or a Balun may be used to convert a singleended signal 3406 to a differential signal or vise versa 3410.

The receiver block 3402 which contains an exemplary embodiment of theinvention next converts the differential radio frequency signal 3410 toa differential intermediate frequency (IF) 3412. Equivalently, singleended signals, or a mixture of differential and single ended signals areutilized in the receiver block 3402.

A large gain range high linearity, low noise MOS variable gain amplifier(“VGA”)3403 is present to adjust the IF signal level 3412. A controlvoltage 3407 controls the gain of the IF signal such that a linearcontrol voltage verses gain response is produced. A linearizationcircuit 3405 produces the linear control voltage from the control signalinput 4302. The IF signal 3412 is next converted down to DC anddemodulated into a base band signal 3414 by a demodulator 3416. At thispoint the base band signal 3414 is suitable for presentation to thevideo input of a television receiver, the audio inputs to a stereo, aset top box, or other such circuitry that converts the base band signalinto the intended information output.

The communication system described is contemplated to provide thefunction described above in one or more circuit assemblies, integratedcircuits or a mixture of these implementations. In particular, the RFfront end 3408 may be integrated in a single chip with receiver 3402.Alternatively, the front end and receiver may be implemented asindividual integrated circuits, on any suitable material such as CMOS.

In addition, the receiving system described utilizes additionalexemplary embodiments that incorporate one or more transmitters and oneor more receivers to form a “transceiver” or “multiband transceiver.”The transceiver contemplated may transmit and receive on differingfrequencies or the same frequency with appropriate diplexer, transmitreceive switching or functionally equivalent circuitry.

The frequency bands and modulation described in the specification areexemplary with the inventions not being limited in scope to anyparticular frequency band or modulation type.

Receiver Front End-Programable Attenuator and LNA

To achieve a low noise figure what is left out of the circuit is oftenas important as what is included in it to achieve a low noise figure. Acircuit containing few components in desirable since each component in acircuit adds to noise generated in the circuit. Switches are oftenincluded early in a signal path to switch in attenuator sections,reducing the level of a signal present. The reduction in signal level isnecessary to prevent a following receiver circuit from being over driveninto distortion.

In an embodiment a large gain range, high linearity, low noise MOS VGA3403 is used as an automatic gain control (“AGC”) amplifier.Additionally, the circuit described as a front end circuit may also beemployed as an AGC amplifier. The AGC amplifier may advantageously beused at any point in the signal processing chain where an adjustablegain and adjustable attenuation according to an external control signalis desired.

In one specific embodiment, a control signal 4302 from an external pinon the integrated circuit is applied to RF front end 3408 and an IF AGCamplifier 3404. The control signal applied to the IF AGC amplifier 3403is first conditioned by a linearization circuit 3405 so that a linearcontrol of the IF AGC amplifier's gain is produced by varying thecontrol signal 4302. The signal output by the linearization circuit 3405is a control voltage 3407.

By way of example, control signal 4302 could be formed by sampling thesync pulses of the base band television signal and averaging theamplitude of the sync pulses over a period of time.

Advantageously, the present invention has eliminated the need forswitches, reducing a major contributor to increased noise figure. In anintegrated switchless programmable attenuator and low noise amplifieronly two elements are present in the signal path to contribute to thenoise figure. First an attenuator is present in the circuit path. Thenext element in series with the attenuator in the signal path is adifferential pair low noise (LNA) amplifier. In the differential pairnoise figure is lowered by introducing a sufficient bias current toincrease a transconductance g_(m) associated with the amplifier. Theincreased g_(m) decreases the noise contribution of the differentialpair.

By eliminating the need for switches it is possible to integrate theprogrammable attenuator and LNA onto a single CMOS integrated circuit.An additional advantage can be realized in using an integratedprogrammable attenuator and LNA as a “front end” of an integratedreceiver. A single integrated circuit can be economically fabricated onCMOS that contains an entire tuner circuit including the front end andthe tuner. Alternatively, the front end and tuner circuits may be onseparate interconnected substrates.

FIG. 35 is an illustration of the input and output characteristics of anintegrated switchless programmable attenuator and low noise amplifier3502. Attenuator/amplifier 3502 simulates a continuously variablepotentiometer that feed a linear amplifier. As the potentiometer settingchanges the signal level at the input to the amplifier changes, and theoutput of the amplifier changes accordingly. The exemplary embodiment isa two radio frequency (RF) port device—the input port 3504 is configuredto receive a single ended input signal from a source 3508 and the outputport 3506 is configured to present a differential signal. In the singleended input configuration one terminal upon which a signal is carried isabove ground reference 3510. In the differential output configurationthe signal is divided and carried on two terminals above groundreference 3510.

In the exemplary embodiment multiple control signals 3512 are applied tothe integrated switchless attenuator and LNA 3502. For example thesesignals are used to program the attenuator to various levels ofattenuation, and for an output smoothness control.

In the exemplary embodiment the differential output 3506 advantageouslytends to provide noise rejection. In a differential outputconfiguration, the signal at one terminal is 180° out of phase from thesignal at the other terminal and both signals are of substantially equalamplitude. Differential signals have the advantage that noise that isinjected on either terminal tends to be canceled when the signal isconverted back to a single ended signal. Such common mode noise istypically of equal amplitude on each pin and is typically caused byradiation into the circuit from external sources, or it is oftengenerated in the circuit substrate itself. Advantageously, the presentinvention uses differential signal transmission at its output. It shouldbe noted that in alternate embodiments of the invention, that a signalended output can be produced from the differential signal by varioustechniques known in the art. Also, equivalently a differential input maybe substituted for the single ended input shown.

FIG. 36 is a functional block diagram of the integrated switchlessprogrammable attenuator and low noise amplifier circuit. This embodimentillustrates how it is possible to eliminate switches that would berequired in a conventional attenuator and LNA.

A resistive attenuator 3601 is configured as a ladder circuit made up ofresistors configured as multiple pi sections 3602. A method of selectingresistor values such that a constant impedance is presented to thesignal source is accomplished as is conventionally known in the art. Anexemplary embodiment utilizes an R/2R configuration. Each pi section3602 of the attenuator 3601 is connected to one input to a differentialpair amplifier 3603. The other input to amplifier 3603 is grounded. Theresulting attenuation produced at the output 3604 is controlled by thenumber of differential amplifier stages that are turned on and thedegree to which they are turned on.

Individual amplifiers 3603 are turned on or off by tail-currentgenerators 3605 associated with each stage 3603, respectively.Generation of the tail currents is discussed in more detail below inconnection with FIGS. 44a and 44 b. In FIG. 36 a zero or one is used toindicate if the corresponding tail-current generator 3605 is turned onor off, that is whether or not a tail-current is present. For example, azero is used to show that no tail-current is present and thecorresponding generator 3605 is turned off. A one represents atail-current generator 3605 that is turned on rendering thecorresponding amplifier 3603 functional. The zeroes or ones are providedby the control lines 3512 of FIG. 35 in a manner described in moredetail in FIG. 43. All of the individual amplifier outputs 3506 aredifferential. Differential outputs 3506 are tied in parallel with eachother. The resulting output 3604 is the parallel combination of the oneor more amplifiers 3608, 3610, 3612 that are turned on. In an exemplaryembodiment of the circuit 55 amplifiers have been implemented, withvarious combinations turned on successively. By using tail currents toselectively turn amplifiers 3603 on and off, the use of switches isavoided.

In this configuration any combination of amplifiers 3603 could be turnedon or off to achieve a given attenuation before amplification of thesignal. However, in a exemplary embodiment of the circuit, adjacentpairs of amplifiers are turned on and off. Groupings of amplifiers inthe on state can be of any number. In an embodiment ten contiguousamplifiers are turned on. The attenuation is adjusted up or down byturning an amplifier tail current off at one end of a chain ofamplifiers, and on at the other to move the attenuation in the desireddirection. The exemplary circuit is controlled such that a group ofamplifiers that are turned on slides up and down the chain according tothe control signals 3512 of FIG. 35.

Any number of amplifiers 3603 can be grouped together to achieve thedesired resolution in attenuation. By using the sliding configuration,input signals 3614 that are presented to attenuator pi sections 3602whose amplifiers are not turned on do not contribute to the outputsignal 3604. It can be seen from FIG. 36 that the signal strength of theoutput is dependent upon where the grouping of generators 3605 areturned on.

FIG. 37 is a simplified diagram showing the connection 3702 of multipleattenuator sections 3602 to the output 3604. An attenuator 3601 is madeup of multiple pi sections 3602 cascaded together. Each pi sectionconsists of two resistances of 2R shunted to ground, with a resistor ofvalue R connected between the non grounded nodes. Tap points 3702 areavailable at the nodes of the resistor R. In FIG. 37 the first set ofnodes available for tap points in the first pi section would be nodes3706 and 3708. After cascading all of the pi sections to form a laddernetwork, a variety of tap points are available, these are noted as nodenumbers 3706-37150 in FIG. 37. A path from the input 3614 to any of thetap points, or nodes on the ladder network yields a known value ofattenuation at the output 3604. If multiple tap points aresimultaneously connected to the attenuator, the resulting attenuation isthe parallel combination of each connection. The combined or averageattenuation at the output terminal can be calculated mathematically or,it can be determined using circuit simulation techniques available incomputer analysis programs.

In addition it can be seen from FIG. 37 that by providing multiple tappoints on a ladder network that in effect a sliding multiple contactaction can be implemented contacting a fixed number of contacts, for anygiven position of the simulated slide 3716. The slide 3716 isimplemented electronically in the embodiments of the invention Theaverage attenuation by contacting a fixed number of these tap points3706-3715 will increase as the slide or switch is moved from the left tothe right on the ladder network. For example, minimum attenuation willbe present when the slider 3716 contacts the force tap points 3706,3707, 3708, 3709 at the far left of the ladder network 3601. The maximumattenuation will be achieved when the slider 3716 is positioned tocontact tap points 3712, 3713, 3714, 3715 at the far right of thenetwork. In the exemplary embodiment 4, contacts are shown, however, inpractice any number of contacts may be utilized.

Mechanical switches are noisy. Mechanical switches are also unreliableand difficult to integrate on a semiconductor device. Returning to FIG.36, in order to be able to integrate a switching function, and toeliminate mechanical parts, a predetermined number of attenuator tapsare switched to the output by using tail current switching ofdifferential amplifiers 3603,3605. The differential amplifiers have theadvantage of being able to be switched electronically with low noise andreliability. The differential amplifiers also provide the opportunity tointroduce a gain into the circuit thereby increasing the signal strengthavailable at the output to produce a low noise amplification. The gainachieved depends upon the number of amplifiers switched in. By changingthe values of resistance in the ladder network and also by increasing ordecreasing the number of amplifier stages that are turned on, theresolution of the attenuator can be varied to suit the needs of thesystem that an integrated switchless programmable gain attenuator andLNA is used in.

FIG. 38 is an illustration of an exemplary embodiment showing how theattenuator 3601 can be removed from the circuit, so that only the LNAsor differential stages 3605 are connected. Reference numerals 3801 to3816 each represent a differential amplifier 3603 and a generator 3605in FIG. 36. In the 0 dB attenuation case shown the signal strength ofthe output would be equal to the gain of the parallel combination of thefour amplifiers that are turned on 3801, 3802, 3803, 3804. The fouractivated amplifiers are indicated by a “1” placed on the circuitdiagram. In an exemplary embodiment in which the sliding tap arrangementis used such that a given number of amplifiers are always turned on theconfiguration of FIG. 38 is necessary such that zero decibels ofattenuation can be achieved when the required number of amplifiers arealways turned on.

In an exemplary embodiment according to FIG. 38, a full 14 dB gain froma combination of ten amplifiers is seen when a ten tap configuration isused with the top set to the 0 dB attenuation position. As theattenuation is “clicked” so that one amplifier at a time is switched, a1 dB per pi section attenuator is placed in series with an amplifier, afull 1 dB of attenuation is not seen/click. In a graph of the controlvoltage versus attenuation curve this would be seen as a change in slopeafter the tenth amplifier is switched in. After the 10th amplifier isswitched in the curve will show a 1 dB/adjustment step.

FIG. 39 shows an exemplary attenuator circuit used to achieve 1 dB/stepattenuation. Each resistive pi section 3602 makes up one step. Thecharacteristic impedance of the embodiment shown is 130 ohms. Usingcalculation methods well known in the art of attenuator design a pi padhaving a characteristic impedance of 130 ohms may be realized utilizingseries resistors R_(s) of 14 ohms or parallel or shunt resistors of1,300 ohms R_(p).

FIG. 40 illustrates an exemplary embodiment of an attenuator forachieving a finer resolution in attenuation. In this embodiment aresolution of 0.04 dB/tap is achieved. In the embodiment shown eachseries resistor R_(s), connected between the shunt resistors in theladder network has a string of series resistors connected in parallelwith it. Each interconnection point between the added resistors 3402provides a tap point that provides a finer adjustment in attenuationvalues.

In implementing an integrated, switchless, programmable attenuator andlow noise amplifier, calculating the overall gain of a parallelcombination of amplified and attenuated signals is analytically complexto calculate. For example, consider an embodiment utilizing 10differential pair amplifiers in the output, connected to 10 differenttap points. Ten signals receiving varying attenuations are fed intoindividual differential pair amplifiers. Gain of the amplifiers variesaccording to an adjustment for monotonicity. The amplified signals arethen combined in parallel to yield the output signal.

Tail currents in the differential output amplifiers are not all equal.The tail currents determine the gain of a differential pair, and areadjusted to provide a specific degree of monotonicity. Thus, the gain ofeach of the differential pair amplifiers varies across the 10interconnected amplifier. The attenuation varies since each tap is takenat a different point to be fed into each of the differential amplifiers.In such an arrangement it would be expected that the middle signal linewould represent the average, yielding an approximate figure for theattenuation and gain of the combination of 10 signal lines. However,this is not the result. Through the use of computer simulation thebehavior of this network has been simulated. In simulating behavior ofthis network it is found that the first tap predominates in defining aresponse from the sum of the 10 taps. The first tap has the leastattenuation and this yields the predominant signal characteristics.

In an embodiment utilizing 10 sliding taps the amplifier gain is aconstant 14 dB. The attenuator range is from 0-25 dB in 1 dB steps. Thisyields an overall range of −11 dB to +14 dB for the combination ofattenuator and amplifiers.

FIG. 41 illustrates the construction of the series and parallelresistors used an integrated attenuator. In this embodiment all of theresistors used are 130 ohms. This is done to control the repeatabilityof the resistor values during fabrication. Ten of these resistors areconnected in parallel to yield the 13 ohm resistor used as the seriesattenuator element R_(s) of FIG. 39. Ten of these 130 ohm resistors areconnected in series to yield 1,300 ohms to realize the parallelresistance legs R_(p) of FIG. 39 of the attenuator. Building theattenuator from unit resistors of 130 ohms also, provides improvedmatching. By matching resistor values in this method variability isminimized to that of the interconnections between the resistors. Thisallows the ratio between series and parallel resistances to remainconstant from pi section to pi section 3602 in the ladder network thatmakes up the attenuator 3601 of FIG. 36.

FIG. 42 is an illustration of an exemplary embodiment utilized to turnon each of the differential amplifiers. This arrangement produces amonotonically increasing output verses control voltage 4202. In thisillustration, five amplifiers 4204-4208 grouped together make up theelectronically sliding tap arrangement. Numbers on the illustrationindicate the fractions of tail-currents relative to the full value usedto turn on each amplifier. Amplifiers are partially turned on at theends of the group. Gradual turn on of the amplifiers at the ends of thegroup is done to control overshoots and undershoots in the amplifiergain. These over shoots and under shoots are seen upon the applicationof a control voltage applied.

Varying a smoothness control provided in a programmable attenuator andLNA to one extreme yields good linearity in the frequency response butovershoots in gain with increases in control voltage. Varying thesmoothness control to the other extreme yields a very smooth gain versescontrol voltage curve with more nonlinearity. The optimum value for thesmoothness control yields a value of monotonicity that is the maximumthat the system can tolerate in the form of data loss throughout thecircuit.

If all five amplifiers of FIG. 42 were turned on with the full value oftail-currents, the gain versus control voltage curve would be as shownin the solid line 4210. By not fully turning on some of the differentialpair amplifiers the overshoot and undershoot in the gain versus controlvoltage curve may be minimized. With the tail-currents configured on thesliding tap as shown in FIG. 42, the gain versus control voltage curvewill appear as shown by the dotted line 4202. In this configuration, themiddle three amplifiers have their tail-currents fully turned on withthe remaining two amplifiers at the beginning and end of the chain onlyhaving their tail-currents half turned on. Equivalently, other weighingof total currents may be used to achieve substantially the same effect.

A plot of gain versus control voltage for the entire integratedswitchless programmable attenuator and low noise amplifier wouldpreferably appear as a staircase over the entire control voltage range.By controlling the turn on of the tail-current, the non-monotonicity ofthe gain versus the control voltage curve is reduced so that the gainmonotonically increases with the application of an increasing controlvoltage to yield the desired stair step shape response, where FIG. 42illustrates one “step” 4202 in the response. Non-monotonicity in gainversus control voltage is not a time dependent phenomenon. The shape ofthe curve tends to depends on the physical implementation of a circuitand a switching arrangement for turning tail-currents on and off.

Non-monotonicity is an undesirable characteristic tends to degradeoverall systems performance. In receiving QAM data the degradation isseen as a loss in received data. By improving the monotonicitycharacteristic of an amplifier linearity of the amplifier is degraded.Gradual switching of the tail-currents causes some differential pairs toonly partially turn on. Differential pairs that are partially turned onintroduce more nonlinearities into the circuit output than a fullyturned on differential pair.

A transistor that is only partially turned on is only capable ofhandling a smaller signal than one that is more fully turned on. Atransistor that is only partially turned on receiving a large inputsignal over drives the transistor producing a distorted output. Thus, bygradually turning on the tail-currents in some of the differential pairamplifiers, the linearity tends to be degraded, however, thisdegradation in linearity allows a monotonically increasing gain versuscontrol voltage curve to be achieved.

Monotonic increase of gain versus control voltage tends to improvesystem performance. In the case of the QAM television signal beingtransmitted through the amplifier a view of a QAM constellation wouldactually be seen to wiggle with tail-currents of all differential pairamplifiers simultaneously and fully turned on. With gradual tail-currentswitching, the constellation is not seen to wiggle, and data is notlost. The problem with the non-monotonicity causing the constellation towiggle is that each time an attenuator value is switched into thecircuit QAM data tends to be lost, thus degrading overall systemperformance of the signal transmitted through the circuit.

As part of an exemplary embodiment's operation, an automatic gaincontrol (AGC) 3512 of FIG. 35 would be generated as one of the controlsignals by external receiver circuitry to adjust the input signal levelpresented to the receiver. This AGC control voltage would be fed into acontrol voltage input 3512 to select a value of attenuation through thecircuit assembly. It is desirable to switch the attenuator such thatwhen the attenuation is adjusted, the data is not lost due to theswitching period. In an exemplary embodiment of the present invention itis necessary to switch a maximum of 0.04 dB per step in attenuationvalue.

FIG. 43 is an illustration of an embodiment showing how individualcontrol signals 4301 used to turn on individual differential pairamplifiers are generated from a single control signal 4302. There aremany ways to generate control signals to turn on the differential pairamplifiers, individual control lines may be utilized, or a digital toanalog converter may be used to transform a digital address to an analogcontrol voltage.

In the embodiment of FIG. 43 to generate the control signals resistors4304 are connected in series between a power supply voltage and groundto create a series of reference voltages at each interconnecting node.The voltages at each node between the resistors is the reference inputfor one of a series of comparators 4306. The reference input of thecomparator connects to a node providing the reference voltage setting.The other input of the comparator is connected to the control voltage4302. When the value of the control voltage exceeds that of thereference voltage for a given comparator the comparator goes from a zerostate to a one state at its output. The zero state is typically zerovolts and the one state is typically some voltage above zero. Thevoltage generated to produce the logic one state is such that whenapplied to a gate of a transistor making up the current tail 4308 it issufficient to turn on the differential pair of amplifiers thatconstitute the low noise amplifier (LNA) controlled by that currenttail.

As can be seen from FIG. 43, all the LNA amplifiers set to be activatedwith a control voltage of the current setting will be turned on. In thisarrangement simply increasing the control voltage simply turns on moreLNA amplifier stages. Additional circuitry is required to deactivatepreviously activated amplifiers such that only a fixed number ofamplifiers remain turned on as the control voltage increases. This isdone so that the sliding potentiometer function can be implemented withthis circuit.

FIGS. 44a and 44 b illustrate an embodiment of one of the individualcomparator stages 4308 of FIG. 43 used to turn on or off individual LNAamplifier stages. In the integrated switchless programmable attenuatorand low noise amplifier the circuitry used to activate individual cellsis duplicated at each attenuator's tap point and interconnected so thata sliding tap can be simulated using a single control voltage, V_(ctr)4302. In describing a cell's operation it is convenient to start withthe control voltage 4302 that is being applied to achieve a givenattenuation value.

To illustrate the comparators operation, a control voltage is applied toeach of a series of comparators, as is shown in FIG. 43. The circuit ofFIGS. 44a and 44 b makes up one of these comparators. FIGS. 44a and 44 bshow the control voltage as V_(ctr), and the reference voltage asV_(ref). These voltages are applied to the gates of a differential pairof transistors (Q1 Q2). The circuit in FIGS. 44a and 44 b surrounding Q1and Q2 functions as a comparator with low gain. The gain of thecomparator is kept low to control the speed of switching on and off thetail-currents of the low noise amplifiers.

In FIGS. 44a and 44 b when the control voltage input V_(ctr) passes thereference level set at V_(ref) the amplifier with its reference setclosest to, but less than V_(ctr) remains deactivated. (The n+1amplifiers where V_(ctr) has not exceeded V_(ref) remain turned off,until activated by V_(ctr).) First the comparator output “current (celln)” goes high. When “current (cell n)”, which is connected to the gateof Q15, goes high it switches the transistor on. Transistors Q16 and Q17are used to deactivate the adjoining current mirror circuit. Amplifier,Amp_(n) is turned off by shunting current away from the current mirror4402, shutting off the tail current Q15. Thus, the current amplifiercell with a comparator that has just been tripped remains turned off.

Comparator output signal “next (cell n+10)” is the opposite state of“Current (cell n)”. The next 10 cells are turned on by the controlsignal “next (cell n+10)”. These cells have not yet had theircomparators tripped by the control voltage present on their inputs. Thusthe bottom of the sliding tap is pushed up and down by the controlvoltage, V_(ctr). In this state transistors Q16 and Q17 in the next 10cells are not conducting current away from the current mirror. Thisallows the current tails of each amplifier, Q15 to conduct causingamplifier Amp_(n) to be turned on in each of the 10 cells.

Note that as a larger number of cells are grouped together, forsimultaneous turn on, a larger number of differential amplifier cells inthe integrated switchless programmable attenuator and low noiseamplifier are required to achieve the same attenuation range.

Once the control voltage has been exceeded for a given cell, the defaultstate for all the previous amplifiers Amp_(n) is to be turned on, unlessthe cell is deactivated by either Q1 or Q2 being activated.

The signal “previous (from cell n−10)” deactivates amplifier cells whenit is in the high state. This signal is supplied from the previousidentical comparator.

In FIGS. 44a and 44 b, a provision for adjusting the abruptness ofamplifier gain is provided. Transistors Q3 and Q10 are being used asvariable resistors. These variable resistors are used to change the gainof the comparator. Varying the gain of the comparator allows theabruptness in the overall amplifier gain to be controlled. Putting ahigh voltage on “smoothness control” causes the drain of Q5 and Q6 to beshorted together. The gain is reduced and a very gradual transitionbetween states is provided by doing this.

A receiver front end such as previously here is described in more detailin U.S. patent application Ser. No. 09/438,687 filed Nov. 12, 1999(B600:33757) entitled “Integrated Switchless Programmable Attenuator andLow Noise Amplifier” by Klaas Bult and Ramon A. Gomez; based on U.S.Provisional Application No. 60/108,210 filed Nov. 12, 1998 (B600:33587),the subject matter of which is incorporated in its entirety byreference, may be used before the fully integrated tuner architecture.

Receiver Frequency Plan and Frequency Conversion

Returning to FIG. 19 a block diagram illustrating the exemplaryfrequency conversions utilized in the embodiments of the invention. AnRF signal 1906 from 50 MHz to 860 MHz that is made up of a plurality ofCATV channels is mixed 1916 down by a first LO (LO₁) 1912 that rangesfrom 1250 MHz to 2060 MHz, depending upon the channel tuned, to a firstIF signal 1918 that is centered at 1,200 MHz. This 1,200 MHz first IFsignal is passed through a first filter bank 1912 of cascaded band passfilters to remove undesired spurious signals. The first frequencyconversion in the receiver is an up conversion to a first intermediatefrequency 1918 higher than the received RF frequency 1906. The firstintermediate frequency is next mixed 1932 down to a second IF 1922.

A second local oscillator signal at 925 MHz (LO₂) 1904, is used to mix1932 the first IF 1918 down to a second IF 1922 signal centered at 275MHz. A second bank of band pass filters 1934 removes spurious outputsfrom this second IF signal 1922, that have been generated in the firsttwo frequency conversions.

A third frequency conversion 1924, or the second down conversion to thethird IF 1926 is accomplished with a third LO (LO₃) 1930 of 231 MHz. Athird filter 1936 removes any spurious responses created by the thirdfrequency conversion and any remaining spurious responses that haveescaped rejection through the previous two filter banks. This third bandpass filter 1936 may have its response centered at 36 or 44 MHz. A 44MHz IF produced by the 231 MHz LO is used in the United States while a36 MHz IF is used in Europe. The LO₃ is adjusted accordingly to producethe 36 MHz IF. The local oscillator's signals are advantageouslygenerated on chip in the described embodiments. However, in alternativeembodiments the receiver implementation need not necessarily be limitedto on chip frequency generation. In the embodiment shown the second LO1904 is advantageously generated by a narrow band PLL circuit 1910 thatincludes a VCO and a control circuit that tends to keep the VCOcentered.

Local Oscillator Relationship

FIG. 45a is a block diagram illustrating the exemplary generation oflocal oscillator signals utilized in the embodiments of the invention.In the embodiment shown the local oscillator circuitry is disposed upona semiconductor substrate 4503. Equivalently the local oscillatorsignals may be produced by circuitry that is not disposed upon asemiconductor substrate. Other suitable materials are printed circuitboards comprising ceramic, Teflon, glass epoxy, and so on. In theembodiment shown the oscillator circuitry is integrated as a part of atuner integrated circuit on a common substrate. The frequency planutilized in the embodiments utilizes a pure third local oscillatorsignal (LO3) 1930, created by direct synthesis 4502 that falls withinthe band of received signals. The first two local oscillator signals(“LO1”) 1902, (“LO2”) 1904 are generated using indirect synthesistechniques utilizing a pair of phase locked loops 4504,4506.

A third local oscillator (“LO3”) 4502 uses direct synthesis, to dividethe second local oscillator frequency LO2 down to create the third localoscillator signal LO3 1930. The local oscillator signals LO1:1902LO2:1904 LO3:1930 utilize differential signal transmission intransmitting the local oscillator signals to the desired mixers 1916,1932, 1924 of FIG. 19 respectively. In alternative embodiments singleended transmission is utilized to conduct the signals to their intendedlocations.

The indirect synthesis of the first and second LOs utilizes a frequencyreference generated by a 10 MHz crystal oscillator 4508. The 10 MHzcrystal oscillator utilizes the previously disclosed differential signaltransmission and a unique design that advantageously tends to provide anextremely low phase noise reference signal.

The PLLs utilize tuning methods to change frequencies, as required whentuning a desired channel or maintaining a desired frequency once set toa desired frequency. The first local oscillator (LO₁) 1902 is producedby utilizing a method of wide band tuning. The second local oscillator(LO₂) 1904 is produced by narrow band tuning. The embodimentsadvantageously utilize a narrow band tuning circuit and method toachieve frequency lock in the narrow band PLL.

Narrow Band PLL 2 and VCO

FIG. 45b is a block diagram that illustrates the relation of the VCO tothe second LO generation by PLL2. Circuitry to generate the second LOfrequency of 925 MHz 1904 includes a narrow band PLL 4506. A componentof the PLL loop is a voltage controlled oscillator (“VCO”) 4532 thatchanges the second LO frequency in response to a control signal 4533.The VCO also operates under the control of a VCO tuning control circuit4535. The VCO tuning control circuit generates a set of control signals4520 that tend to maintain an optimal range of control voltage in theVCO that in turn tends to provide a valid frequency lock state in thePLL. The VCO tuning control circuit is controlled via external signallines that accept external commands and provide status indications 45104512 4514 4516 4518 that tend to be useful for controlling receiveroperation.

FIG. 45c is a block diagram of an embodiment of a VCO 4532 utilizing atuning control circuit 4535. A control voltage 4533 acts on the VCOcircuit 4532 to produce an output frequency 1904. In the VCO circuit anincreasing control voltage typically produces an increasing outputsignal frequency f_(OUT). The control voltage typically provides a fineresolution in setting the VCO frequency. The fine setting is susceptibleto disruption due to temperature and process variations typical in VCOimplementations. Typically a predetermined control voltage designed tofall near the middle of a VCO's tuning range places the VCO at thecenter of a tuning range. It is desirable to have a VCO that tends tohave a linear relationship between control voltage 4533 and frequencyoutput 1904. However, a linear relationship tends to be difficult tomaintain, especially in an integrated circuit.

In an integrated circuit, process variations and temperature effectstend to work against maintaining the linear relationship. It isdesirable to provide a VCO having performance that tends to be immune tothese effects. A sliding window function that is capable of trackingvariations in circuit performance is provided by a VCO tuning controlcircuit 4535. The sliding function is provided by changing a VCO tankcircuit's resonant frequency by varying its capacitance.

A VCO that tunes linearly at one temperature may fail to maintainlinearity at an elevated temperature. Likewise, a linearly tuning VCOfabricated in one lot run may be found to tune non-linearly whenproduced in a subsequent production run. Temperature and process effectsmay also cause a controlled voltage range to produce a range of outputfrequencies at f_(OUT) that are outside of a desired tuning range. A VCOintegrated onto a semiconductor substrate 4503 tends to require animproved phase noise specification over a particular tuning range.

In an exemplary PLL, with a lock range of 922 MHz to 929 MHz, suitablefor use in a cable tuner disposed on a CMOS integrated circuitsubstrate, a phase noise specification sufficient for NTSC and QAMreception tends to be desirable.

To counteract temperature in process variations in an integrated VCO,the tuning control circuit 4535 is utilized. In an embodiment the tuningcontrol circuit 4535 is disposed upon the same substrate 4503 as anintegrated VCO 4532. In an alternative embodiment the tuning controlcircuit 4535 is implemented off of the substrate.

The tuning control circuit has multiple inputs. It is supplied with a“clock” input 4514 to provide sequencing in performing its internaloperations. In the exemplary embodiment the clock signal is derived fromthe 10 MHz reference signal 4508 of FIG. 45a. An indication of externalcircuitry state 4510 is input to the tuning controls circuit. The“state” signal is derived from the VCO's loop filter. A “reset” line4512 is provided as an input to reset the internal tuning controlscircuitry prior to commencement of a new tuning process cycle.

The tuning control circuit produces an output to the VCO 4532 comprisingone or more (“n”) control lines 4520 that control VCO 4532 tuningcircuitry. Such tuning circuitry may be one or more circuit componentthat sets the VCO tuning range. In an embodiment of the invention sixcontrol lines 4520 are provided.

The tuning control circuit 4535 provides two additional outputs. An “inlock” output 4518 provides an external indication that a phase lockcondition in the VCO has been achieved. The output labeled “done” 4516provides an indication that the tuning control circuit has finishedperforming its function of centering a VCO tuning range.

FIG. 45d is a block diagram of an embodiment of a VCO having a tuningcontrol circuit and showing tuning control circuit interaction withmajor VCO components. A typical VCO as known to those skilled in the artcomprises circuitry that implements the subsystems shown in FIG. 45d.Typical VCO subsystems comprise a gain block 4599, a feedback network4505 and a summing junction 4507 that couples the amplifier output, asmodified by the feedback network, to the amplifier input. Thesefunctions are often implemented by circuit components that possessinterconnections that are not as easily identifiable as shown. However,in any functioning oscillator the functional subsystem andinterconnections as illustrated are present.

A VCO is an oscillator that produces a variable frequency outputf_(out), that is proportional to a control voltage input 4533. A VCO istypically integrated on an integrated circuit substrate 4503. Major VCOcomponents of a VCO comprise an amplifier 4599, a source of feedback,such as feedback network 4505 typically comprising a resonant tankcircuit, and a path to couple the feedback to the amplifier's inputrepresented by a summing junction 4507.

The VCO shown 4532 illustrates in block diagram form the concept thatfor oscillations to be sustained an energy producing element, such asamplifier 4599, provides energy to a feedback network 4505 that byvirtue of its interconnections feeds back a portion of signal f_(OUT)back to the input of amplifier 4599. Feedback is typically provided by adirect connection. However, feedback is also accomplished throughradiation, or a parasitic path, such as through a power supply coupling.To sustain oscillations, the feedback loop must satisfy the Barkhausencriteria at f_(out): G(j2πf_(out))H(j2πf_(out))=−1, whereG(j2πf_(out))is an amplifier transfer function and H(j2πf_(out)) is afeedback network transfer function. If Barkhausen criteria is satisfied,the oscillator will oscillate to produce an output frequency, f_(out)1904.

Feedback network 4505 typically comprises frequency selective elements4509 4511 that form a tuned circuit exhibiting resonance in parallel (asshown in FIG. 45e), series or a combination of series and parallel. Sucha circuit is often referred to as a resonant tank. By varying the tunedcircuit element's value contained in feedback network 4505 the outputfrequency of oscillation f_(out) may be varied. Variation of circuitelement values is accomplished with control voltage 4533 and controllines 4520. The control lines set a frequency tuning range and thecontrol voltage adjusts the frequency within a frequency range setthrough the control lines.

FIG. 45e is a schematic of the feedback network 4505 that allows thefrequency of oscillation to be adjusted. The feedback network comprisescapacitive 4511 and inductive 4509 circuit elements having frequencydependent responses. The feedback network typically comprises multiplecircuit elements to produce an overall frequency response. Equivalentlythe feedback network is intertwined with the amplifier circuit (or gainstage) (4599 of FIG. 45d). For example, a feedback network comprising anLC tank circuit as shown in FIG. 45e will resonate at a frequencydependent upon the combined values of inductance 4509 and capacitance4511. If a variable capacitance 4515 is included, as shown in FIG. 45f,a resonant frequency may be tuned over a range of frequencies byadjusting the capacitance 4515. Alternatively, an inductor 4509 may beof the variable type to adjust the output frequency 1904. However, anadjustable capacitance 4511 is typically easier to fabricate on anintegrated circuit substrate than a tuned inductor 4509.

FIG. 45f is a schematic of a feedback network that allows the frequencyof oscillation to be adjusted continuously by varactor tuning. Varactorstypically provide a fine tuning range of adjustment in a VCO. In anembodiment, a continuously adjustable capacitance is provided byvaractor diodes 4515. A varactor diode is a diode that possesses avarying amount of capacitance. The amount of capacitance depends upon alevel of direct current biasing the varactor diode. To set thevaractor's tuning range, a fixed capacitance 4513 is typically used. Thefixed capacitor typically gets the tuned circuit close to a desiredfrequency, and the varactor fine tunes the desired frequency. In analternate embodiment, a network of discretely switched capacitors may beused in place of fixed capacitor 4513. In the later describedarrangement utilizing discretely switched capacitors, discrete ranges oftunable frequencies, with each range being continuously tunable, areprovided.

With discrete capacitor tuning it is desirable to select the value ofcapacitance by electronically adding or removing a capacitor, withoutmechanical switching. With electronic switching of capacitor values aresonant center frequency for the network is defined by one or morecapacitances that are switched in, combined with the capacitance as setby the varactor's current bias voltage. The capacitance range of thevaractor sets the tuning range of the feedback network.

The varactors in the embodiments of the VCO are fabricated from NMOStransistors 4517. The feedback network 4505 shown provides a tuningrange defined by a series combination of capacitance provided by one ormore varactors 4515 combined in parallel with a fixed capacitor 4513.The varactors provide a capacitance that is variable in response to abiasing control voltage 4533 applied. The varactors are disposed suchthat when a control voltage 4533 is applied to a varactor diode, it isback biased and no current flows. In an embodiment appropriate DCblocking capacitors may be utilized to prevent current flow from thecontrol voltage line 4533.

A varactor is typically constructed as a diode having two leads.However, a discrete device package is incompatible with integratedcircuit construction. In an integrated circuit a varactor may becompactly constructed from an NMOS transistor.

In the embodiments a varactor diode is constructed by shorting a drain(“D”) and a source (“S”) leads (or terminals) of an NMOS transistor4517. The coupled drain and source form one terminal of the varactor,and the gate forms a second terminal of the varactor. By shorting thedrain and source leads of an NMOS device 4517 a bulk resistance 4519from drain to source is present. The bulk resistance is modeled 4519 bya parallel combination of two resistors each of value R. In an NMOStransistor current does not substantially flow from gate (“G”) to eitherof the drain D or source S terminals. Therefore, a separation of chargeor capacitance is created from the first terminal formed by the gate tothe second terminal formed by the shorted drain and source through theparallel combination of two resistors R. A DC voltage applied to theNMOS varactor produces a variable capacitance that is inverselyproportional to the applied DC voltage.

NMOS transistors are a type of MOSFET transistor, which in turn is atype of field effect transistor, or FET. Equivalently, other types ofFETs could be utilized to form a varactor, such as a PMOS device.

FIG. 45g is a graph of capacitance versus control voltage applied to anNMOS varactor. As can be seen from this graph, varactor capacitance 4511tends to be inversely proportional to an applied control voltage 4533. Aportion of the curve tends to be linear 4521. It is desirable to utilizethe linear portion of the tuning curve to tune the VCO. Such a curve isoften referred to as a C-V curve.

FIG. 45h is a graph illustrating average capacitance achievable with anNMOS varactor. Here, a family of various C-V curves are presented fordifferent control, or source voltages.

Equivalent series resistance or ESR is a figure of merit for acapacitor. The ESR of an NMOS varactor is the drain source resistance ofthe shortened leads. In an exemplary design, the NMOS FETs used to formthe varactors have a W/L ratio equal to (20.0/0/35) that is repeated 36times.

V_(S) is the controlled, or source voltage applied to the shorted sourceand drain leads of an NMOS varactor. V_(g) on the horizontal axisrepresents the voltage applied to the gate of an NMOS varactor. As thegate voltage is varied from zero to a maximum voltage, the capacitanceswitches between a depletion capacitance (“C_(dep)”) and an oxidecapacitance (“C_(ox)”). The total charge transferred during each cycleof voltage variation on the gate, such as when a varying noise or RFsignal is present in the circuitry, is a measure of the effectivecapacitance. The effective capacitance is represented by the area underthe C-V curve. Thus, voltage variations in the C-V switch thresholdsmodifies the effective capacitance of an NMOS varactor. Thus, flickernoise in the NMOS device tends to cause frequency modulation of the VCOby changing the capacitance and in turn changing the frequency producedby the VCO.

The capacitance produced from an NMOS connected to form a varactor is anaverage value of the device's capacitance. When the NMOS' applied gateto source voltage (“V_(gs)”) is less than an inherent threshold voltage(“V_(t)”) of an NMOS transistor, the transistor is in the “off state”and has a capacitance equal to a depletion capacitance (“C_(dep)”) ofthe NMOS. This is a relatively small value of capacitance.

When V_(gs) exceeds V_(t), the NMOS is in an “inverted state” where agreater oxide capacitance (“C_(ox)”) is produced. A changing gatevoltage produces a capacitance that is not linear, but rather an averagecapacitance. The capacitance switches between a low capacitance and ahigh capacitance value depending upon signal swing present across theNMOS, such as is present in an RF signal.

The value of average capacitance depends upon the time the MOSFET is“inverted” compared to the time that it is “off”. The voltage gating thevaractor on and off is the voltage swing across the varactor. Forexample the voltage swing across the varactor is the result of the VCOoutput's RF signal swing being present across the varactor. Effectivecapacitance depends upon a charge transfer which is equal to the areaunderneath the CV curve. Thus, an integration of the area under the CVcurve for a given voltage swing (“V_(g)”) represents the effectivecapacitance obtained.

Further, this average capacitance is a linear function of the signalswing and the control voltage. As the voltage on the source drainconnection (“V_(s)”), which is the control node, is changed, theswitching point is changed, since the voltage on the gate V_(g) mustexceed the voltage on the control node by V_(t) before the large oxidecapacitance is formed. Thus, by changing the control voltage V_(s), thecapacitance of the NMOS varactor is changed.

FIG. 45i is a schematic of an embodiment of a VCO 4532 that includes anamplifier 4599, a feedback network 4505 and summing function 4507 in itscircuitry. The embodiment shown utilizes NMOS varactors 4517 to providefrequency control.

The amplifier circuit 4599 consists of a pair of NMOS driver transistorsM1 M2. The NMOS drivers each possess an inherent capacitance C_(gs) thattends to contribute to the tuning of the VCO.

Transistor M1 has its source coupled to ground. The drain of M1 iscoupled to the gate of M2, a first terminal of a first inductor 4509,the first terminal of a first varactor 4515 and a set of first terminalsof a first bank of six capacitors 4528. A set of second terminals of thefirst bank of six capacitors are each coupled to one of a first set ofsix transistor switches 4527 drains. The sources of the switchingtransistors are coupled to ground. The gates of each of the switchingtransistors are coupled to individual control lines b₁ through b_(n)4520 that make up the n control lines that originate from the tuningcontrol circuit (4535 of FIG. 45d).

Transistor M2 has its source coupled to ground. The drain of M2 iscoupled to the gate of M1, a first terminal of a second inductor 4509,the first terminal of a second varactor 4515 and a set of firstterminals of a second bank of six capacitors 4528. A set of secondterminals of the second bank of six capacitors are each coupled to oneof a second set of six transistor switches 4527 drains. The sources ofthe second set of switching transistors are coupled to ground. The gatesof each of the switching transistors are coupled to individual controllines b₁ through b_(n) 4520 that emanate from the tuning control circuit(4535 of FIG. 45d).

The second terminals of the first and second varactors are coupledtogether and to the control voltage 4533 supplied by the tuning controlcircuit (4535 of FIG. 45d). The second terminals of the first and secondinductors are each coupled to the source of transistor M3 of theadaptive bias circuit 4522.

The adaptive bias circuit 4522 comprises a PMOS transistor M3 with itsdrain coupled to a voltage supply V_(DD) and a first terminal of acapacitor 4531. The second terminal of capacitor 4531 is coupled to thegate of M3. The gate of M3 is also coupled to the first terminal of aresistor 4524. The second terminal of resistor 4524 is coupled to theadaptive bias control line 4530 that is supplied by a constant G_(m)bias cell 4536.

Adaptive bias causes the transconductance of transistors M1 and M2 toremain fixed. Adaptive bias 4522 is provided by a PMOS transistor M3that tracks temperature and process variations by virtue of beingfabricated by common IC processing. Variations in process andtemperature create a varying voltage at the gate of PMOS transistor M3.

The adaptive bias control line 4530 is coupled to the gates oftransistors M4 and M5 in the constant G_(m) bias cell 4536. The constantG_(m) bias cell is representative of the functions needed to implementadaptive bias and is conventionally constructed as is known to thoseskilled in the art. The constant G_(m) bias cell tends to maintain thetransconductance of M6 (g_(m)) at a value of 1/R2 through localfeedback. Current I varies with temperature and process to ensure this.The value of R2 is scaled through an amplifier gain. Appropriate scalingof M1 and M2 with respect to M6, and of M3 to M5 gives ag_(mM1/M2)=k(1/R2)=k(g_(mM3)). Thus, a constant g_(m) tends to bemaintained in transistors M1 and M2.

In the constant G_(m) bias cell the drains of M5 and M4 are coupled toV_(DD). The gate of M5 is coupled to the source of M5. The source of M5is also coupled to the drain of M7. The source of M7 is coupled to afirst terminal of R2. The second terminal of R2 is coupled to ground.The source of M4 is coupled to the drain of M6 and the gate of M6. Thesource of M6 is coupled to ground.

Maintaining a constant transconductance in M1 and M2 assists inmaintaining a sliding window. The sliding window that is beingmaintained is the upper and lower limits of the VCO control voltagerange. For the transconductance of M1 and M2 to remain constant, theirV_(gs) must move in response to temperature and process variations. AsV_(gs) moves, it is desired to have the window move to track thischange. The capacitance obtained across the varactor is dependent uponthe V_(gs) of M1 and M2. Thus, if the V_(gs) of M1 and M2 changes, it isdesirable to have the window change in a manner responsive to the changeof the V_(gs) of M1 and M2.

FIG. 45j is a schematic of an equivalent circuit model of the VCO ofFIG. 45i. In an embodiment, a design provides specific phase noiseperformance. The noise contribution is primarily due to flicker noise ofthe transistors, varactors, and the bias circuit.

In modeling an equivalent circuit, as long as the tunable capacitance isa small fraction of the fixed tank capacitance, the flicker noise(“1/f”) contribution of the varactors is minimal.

Up-conversion of 1/f noise is minimized by maximizing the gate thresholdvoltage (“V_(gt)”) of M1 and M2 of FIG. 45i, and making the transistorsreasonably large. In making the transistors large the total gatecapacitance present in the circuit is a constraint. The biasingtransistor M3 of FIG. 45i is made wide and short to maximize gate areaand minimize its head room impact. Headroom impact refers to the factthat to reduce power consumption the inductors 4509 are not coupleddirectly to V_(DD). As the W/L ratio of M3 is reduced, a larger voltageis dropped across the drain and source terminals of M3. To providesufficient headroom in the described embodiment it is desired tomaintain V_(DS)>(V_(SG)−V_(t)). In a final effort to reduce 1/f noise,the gate of transistor M3 of FIG. 45i is filtered by a 100 k OHM on-chipresistor 4524 and a 0.1 μF external capacitor 4531, both of FIG. 45i.The filtering ensures that noise from the small bias devices does notadversely affect the overall noise performance of the VCO core shown inFIG. 45i. The low pass filter possesses a 10 ms time constant that doesnot affect startup as the external 0.1 μF capacitor is initially chargedthrough a switch having a worse case on-resistance of substantially 50OHMS. The 0.1 μF capacitor and 50 OHM resistance provide an acceptabletime constant for circuit performance.

The small signal circuit model shown in FIG. 45j is a reasonableapproximation of the VCO since the switched capacitors and varactors aredesigned to have a Q that is much greater than the inductors. Thermalnoise arising from the substrate and gate resistance is minimizedthrough careful design and layout techniques known to those skilled inthe art. The equivalent parallel resistance of the tank is 2R, where Ris approximately equal to (Q²)r.

FIG. 45k is a schematic of a tuning control circuit controlling switchedcapacitors tending to center a varactor tuning range. In FIG. 45k,variable capacitors 4511 of FIG. 45f are represented by a single fixedcapacitor 4513, a series of switched capacitors C₁ 4528 through C_(n)4528, and a continuously variable capacitance provided by a pair ofvaractors 4515. The capacitors utilized in the circuit may be of anytype including those suitable for integrated circuit fabrication. In anembodiment, metal fringe capacitors are used for the switchedcapacitors. The parallel combination of the capacitors provides therequired overall capacitance C as shown in FIG. 45e. In alternativeembodiments, capacitance in the tuned circuit may be made up of anynumber or combination of fixed capacitors and switched capacitors.Capacitors C₁ through C_(n) are discrete capacitors that are added orremoved from the tuned circuit by a field effect transistor (“FET”)switch.

Each switch is activated through an individual control line that is partof a bus of control signals 4520 emanating from the tuning controlcircuit 4535. In alternative embodiments, the number of control linesmay be reduced to less than one per switch by addressing a demultiplexerthrough one or more multiplexed lines. The presence of a voltage on anyone of the given control lines, sufficient to turn on the channel of thefield effect transistor, effectively couples the capacitors 4528 to thetunable resonance circuit 4505.

FIG. 46a is a schematic of a PLL having its VCO controlled by anembodiment of the VCO tuning control circuit. A VCO tuning controlcircuit 4535 is provided to tune a VCO 4532 that is contained in anexemplary narrow band PLL 4506 that generates an I and a Q 925 MHz localoscillator signal 1904. In the embodiment shown the local oscillatorsignal is a differential signal. However, in alternate embodiments asingle ended signal is equivalently utilized.

The tuning control circuit makes use of a temperature and processdependent moving window of acceptable control voltages defined by arange of voltages that vary with temperature and process. The movingwindow tends to aid in optimally choosing a range of valid controlvoltages for the PLL that tend to aid in attaining a frequency lock. Thecontrol circuit uses the moving window to center a varactor diode's(4515 of FIG. 45k) tuning range by adding or removing capacitance.Centering tends to avoid gross varactor non-linearities by causing arange of control voltage being utilized to fall on a linear operatingregion of a C-V curve. Also, the circuit tends to mitigate dead bandconditions and tends to improve loop stability over process andtemperature conditions.

Process and temperature variations cause variations in VCO performance.Process variations refer to inconsistencies in the manufacturing processthat can result in wafer-to-wafer and/or chip-to-chip differences. A VCOintegrated on a chip can be up to ±20% off in its frequency range.Environmental effects primarily consist of temperature. Pressure andhumidity can have a second order effect on performance. Immediatecalibration at power up is done to center the varactor diodes at themiddle of their tuning range. This is done by switching in capacitorsand monitoring loop voltage. To center the VCO's frequency tuning rangethat is provided by the variable capacitance of the varactors, theembodiments of the invention immediately calibrate the VCO by adding orremoving capacitance. Switching capacitors in or out of the circuitcenters the varactor's capacitance range at the middle of the VCO'stuning range. To monitor centering of the varactors a window comparatoris used to evaluate the state of a control voltage that is used to tunethe VCO. The window comparator determines when the control voltage iswithin the VCO's preferred control voltage range to improve the PLLperformance.

The VCO tuning control circuitry 4535 controls the VCO 4532 of aconventional PLL 4506. The PLL is conventionally constructed as is shownin FIGS. 17-18. A reference divider 4610 is controlled by externallysupplied frequency select lines 4608. The PLL comprises a crystaloscillator 4606 that inputs a stable frequency to the programmablereference divider 4610. In the embodiment shown the crystal oscillatoris constructed as shown in FIGS. 7-16. In alternative embodiments thecrystal oscillator is conventionally constructed as is known by thoseskilled in the art. The reference divider is conventionally constructedas is known by those skilled in the art. The reference divider in turnoutputs a frequency 4612 that is based upon the reference frequency to afirst input of a phase detector 4614. The phase detector isconventionally constructed as is known by those skilled in the art. Asecond input 4616 to the phase detector is a current output of the VCO4532.

FIG. 46b illustrates a pulse train output of the phase detector. A pulsetrain 4620 is derived from the VCO output signal 4616 and the referenceoscillator signal 4608 as shown.

Returning to FIG. 46a, the phases of the two phase detector inputs 4612,4616 are compared in the phase detector 4614. A pulse train representingthe phase difference is output 4620 from the phase detector and coupledto the input of a charge pump 4622. The charge pump is conventionallyconstructed as is known by those skilled in the art. The output of thecharge pump is fed into a low pass filter 4624. The output of the lowpass filter 4624 is fed into the control voltage input of the VCO 4532.The VCO outputs an image and quadrature signal 1904 at a frequency asset by the frequency select line 4608.

The voltage controlled oscillator 4532 is conventionally constructed,and comprises a variable capacitance used to tune the output frequency.VCO 4532 additionally comprises a series of switchable capacitorsutilized to center the tuning range of the variable capacitance elementscomprising the VCO. The switchable capacitors are controlled by signalsemanating from the VCO tuning control circuitry 4535. The controlsignals 4520 are routed from tuning register 4630 to the VCO 4532.

The VCO tuning control circuitry 4535 utilizes a control signal called“state” 4510 taken from low pass filter 4624. The voltage signal “state”4510 is input to the positive inputs of a first LSB comparator 4634 andthe positive input of a second MSB comparator 4636. The negative inputsof comparators 4634 and 4636 are coupled to DC reference voltages V1 andV2. These reference voltages shift depending upon temperature andprocess conditions.

Voltages V1 and V2 are taken from a resistive divider circuit thatutilizes a transistor to track process and temperature variations. Aconventional voltage reference 4607 outputting voltage at level V1 isapplied to a first terminal of a first resistor 4603 and the negativeinput of msb comparator 4636. A second terminal of the first resistor iscoupled to a first terminal of a second resistor 4605 at node 4637. Asecond terminal of the second resistor 4605 defining voltage thresholdV2 is coupled to a drain of a transistor M4. The drain of M4 is coupledto the negative terminal of lsb comparator 4634. A source of M4 iscoupled to ground, and a gate of M4 is coupled to node 4637.

Comparator 4634 outputs signal lsb and comparator 4636 output signalmsb. Voltages V1 and V2 set thresholds to form a sliding window whichmonitors the state of the closed PLL by monitoring voltage 4510 at lowpass filter 4624. Control voltage 4510 is taken as the voltage across acapacitor 4613 in the low pass filter that induces a zero in the loopfilter 4624. Thus, the control voltage is a filtered version of thecontrol voltage of the PLL loop, and thus tends to have eliminatedspurious components present on the VCO control line.

Signals msb and lsb are fed in parallel to a 2 input AND gate 4640 and atwo input exclusive NOR gate 4642. The output of exclusive NOR gate 4642is fed into the D input of a DQ flip-flop 4644. The Q output of theflip-flop is fed into a two input AND gate 4646, whose output is in turnfed into the clock input of a 6-bit bi-directional tuning register 4630.Returning to AND gate 4640 its output is fed into the shift left/rightinput port of the 6-bit bi-directional tuning register 4630.

The reset signal 4512 is based on the output of low pass filter (“LPF”)4624 and is applied to the VCO control circuit as described below. Thereset signal 4512 is generated external to the circuit depicted in FIG.46a and is an input to this circuit as shown in FIG. 46a. Low passfilter 4624 takes its input from charge pump 4622's output. A firstshunt capacitor 4609 has a first terminal coupled to the LPF input andit has a second terminal that is shunted to ground. Resistor 4611 has afirst terminal coupled to the LPF input and a second terminal coupled tothe first terminal of a capacitor 4613. A second terminal of capacitor4613 is coupled to ground, a drain coupled to the first terminal ofcapacitor 4613, and a gate that receives a reset signal 4512 utilizedthroughout the VCO control circuit. The reset signal is coupled to an Rterminal of DQ flip-flop 4644, a reset terminal “R” of the 6-bitbi-directional tuning register 4630, the “R” input of DQ flip-flop 4617,and a first input of a two input OR gate 4619.

Clock signal 4514 is based on the divided reference oscillator signal4612. Division of the reference signal is accomplished in anyconventional manner, known by those skilled in the art. Clock signal4514 is coupled to the clock inputs of DQ flip-flops 4644 and 4617, aclock input of the 6-bit bi-directional tuning register 4630, andin-lock detector 4648. The clock signal is also applied to an invertedsecond input of the two input AND gate 4646.

Threshold voltages V1 and V2 are in fixed relationship to each other butvary in their voltage levels. The pair of voltage thresholds, V1 and V2,utilize a MOSFET transistor M4 4635 to provide a sliding windowfunction. The window is formed by the voltages V1 and V2. The actuallocation of the window is set by the V_(gs) of MOSFET M4 since thetemperature and process changes present in M4 cause the value of V1 andV2 to change. However, the difference in voltage between V1 and V2remains constant.

For example, for a change in temperature, the V_(gs) of M1 and M2 wouldchange. A change in the V_(gs) of M1 and M2 causes the capacitance ofthe varactor to change. If a window that did not track the change ofV_(gs) was not provided, then at elevated temperature the loop would notlock. At start-up, when the chip is at room temperature, V1 is set to1.5 volts and V2 is set to 1.0 volt. The phase lock loop will attempt tolock with a voltage between 1.0 and 1.5 volts. Overtime, the chiptemperature increases causing the V_(gs) of M1 and M2 to change. Thecapacitance changes in the varactor cause the VCO to move away from thepreset window. If the PLL tried to acquire lock at the elevatedtemperature, it would not be able to do so within a voltage range of 1.0to 1.5 volts.

MOSFET M4 has the effect of making the voltages at V1 and V2 notabsolute values. However, the difference between V1 and V2 remainsconstant and fixed. At room temperature, V1 and V2 may be 1.5 and 1.0volt, respectively. However at 85° C., they may drift to 2.0 and 1.5volts, respectively. The V_(gs) of M4 changes with the elevatedtemperature. The voltage at the tap point 4637 also increases withtemperature forcing the position of the window defined by V1 and V2 tomove tracking the V_(gs) of M1 and M2.

Narrow Band VCO Tuning

FIG. 47a is a process flow diagram illustrating the process of tuningthe VCO with an embodiment of a VCO control circuit. Initially thecontrol voltage (4510 of FIG. 46a) is evaluated to see if it fallswithin a predetermined window 4702. If the voltage is within the desiredrange, the time it has remained so is determined 4704. The PLL tends tobe in a state of lock when the control voltage applied to the VCO hasremained unchanged for a predetermined period of time. If the voltagedoes not remain in range for the predetermined time, the process isreinitiated by looping back to the beginning. If the control voltageremains in the range for the predetermined time, the loop is deemed inlock, and the process is ended 4712.

Returning to block 4702, if the control voltage is out of range, adecision is made 4706 based on whether the control voltage is above orbelow the desired range. If the control voltage is greater than thecontrol voltage range, a capacitance is removed from the VCO circuit4708. The process flow is routed to the beginning of the process, wherethe control voltage is again reevaluated 4702.

Returning to block 4706, if the control voltage is below the desiredrange a capacitor is added 4710. Next, the process routes the flow backto the beginning of the process where the control voltage is reevaluated4702.

The VCO tuning control circuitry 4535 of FIG. 46a functions to carry outthe process of FIG. 47a. If the voltage of the loop lies outside thewindow defined by the threshold voltages V1 and V2, the clock input tothe 6-bit bi-directional tuning register 4630 is enabled. This registerfunction may be provided by a conventional circuitry known in the art toprovide this function and is not limited to the circuitry depicted. A“lock time out” circuit 4648 of FIG. 46a is reset on the rising edge ofthe clock signal to the 6-bit bi-directional tuning register 4630 ofFIG. 46a. The “lock time out” circuit is conventionally constructed andis not limited to the components depicted in FIG. 46a.

If control voltage 4510 exceeds the upper threshold set by thecomparators, zeros are shifted through the register 4630. A zero voltagedecreases the capacitance in the VCO tuning circuitry by switching out acapacitance controlled by one of the 6 control lines 4520.Alternatively, any suitable number of control lines may be used otherthan the exemplary six. This shifting of values in a register allows oneof six exemplary capacitor switch control lines to be activated ordeactivated, an evaluation made and another line activated ordeactivated so that the previous tuning setting is not lost. Thisfunction may be implemented by passing a value (on or off) down a lineof capacitors by shifting or by activating a capacitor associated with agiven line and then a next capacitor without shifting the capacitancecontrol signal.

If the control voltage 4510 is less than the lower threshold voltage ofthe comparator 4634, ones are shifted through the 6-bit bi-directionaltuning register. The ones increase the capacitance applied in the VCOtuning circuit by switching in a capacitance controlled by one of the 6control lines 4520.

Once control voltage 4510 enters the predetermined valid range ofoperation as set by voltages V1 and V2, the shift register 4630 isdisabled. At this time, the locked time out circuit 4648 is enabled. Ifthe lock time out circuit remains enabled for the predetermined timeperiod, that satisfies the in lock condition for the PLL, and the clockto the DQ flip-flop 4644 is disabled, thus disengaging the controlcircuit. The functions described in this paragraph are constructed fromstandard logic components known to those skilled in the art, and are notlimited to those components depicted in FIG. 46a.

FIG. 47b is a flow diagram of a PLL start up and locking process for anembodiment of the invention.

A PLL start-up process is utilized to ensure that all inputs to the PLLare in the proper initial state and applied at the proper time. ThePLL's start-up and locking process is completed when the PLL achieves asteady state. In the steady state condition, the PLL is set to belocked.

In response to a control signal, the PLL start-up and locking process isinitiated 4701. In an embodiment, a controller utilizes a bus structureto receive data indicative of circuit performance, and to send commandsto a circuit such that the coordination of circuit functions isaccomplished.

After initiation of the process, the logic circuits are reset 4703.Logic signals to be reset comprise a state signal 4510, and a resetsignal 4512 that are inputs to the tuning control circuit 4535 of FIG.46a.

The next process step is directed to setting an initial VCO oscillatorfrequency. A tuning register 4630 (of FIG. 46a) is set to produce anoutput of all ones at process step 4705. An output of all ones causesall capacitors in the VCO to be switched into a feedback network circuit4505 through control lines 4520 (shown in FIG. 45k). With a maximumvalue of capacitance switched into the feedback network, the VCO istuned to its lowest frequency where the frequency F is given by therelation,

F=1/(2π{square root over (LC)})  (7.25)

F=frequency in Hz (hertz)

L=inductance in H (henry)

C=capacitance in F (farad)

Next, the zero cap in a loop filter 4624 (of FIG. 46a) is zeroed in thenext process step 4707.

The tuning control circuit 4535 has now been initialized and VCO tuning4709 is commenced. To tune the VCO, an MSB and an LSB signal are sampledevery 64th clock cycle 4711 in an embodiment. The MSB signal is theoutput of comparator 4636 of FIG. 46a. The LSB signal is the output of acomparator 4634 also shown in FIG. 46a. The action taken in tuning thePLL depends upon the state of the MSB and LSB signals. First, anevaluation is made to determine if the MSB and LSB signals are bothequal to one 4713. If the signals are both in the ones state, acapacitor is switched into the circuit 4720. The state of the circuitcontinues to be monitored and if the MSB and LSB are not equal to one, afurther evaluation is made. Next, the MSB and LSB are evaluated todetermine if both signals are equal to a zero value 4716. If bothsignals are equal to the zero state, a switched capacitor is removed4722. The signal continues to be monitored every 64th clock cycle 4711and when the MSB and LSB signals are not both equal to one or zero, adetermination is made as to whether the MSB signal is equal to zero andthe LSB signal is equal to one 4718. If the signal does not meet thiscondition, the signal continues to be monitored with the capacitanceadjusted until the MSB is equal to zero and the LSB is equal to one forthree clock cycles. Once this condition has been met, the PLL is deemedto be in lock 4724. The circuit condition continues to be monitored andif the PLL remains in lock for 15 reference clock cycles, the tuningcircuit is disabled 4726, and the process is ended 4728.

FIG. 47c is a graph of a family of frequency versus control voltage forvarious capacitor values that illustrates the use of comparatorhysteresis to aid in achieving a frequency lock condition. The firstembodiment of the invention does not utilize hysteresis. An alternativeembodiment of the invention utilizes hysteresis. Comparators 4636, 4634of FIG. 46a are shown as having hystersis incorporated in their design.Returning to FIG. 47c, the comparator's hysteresis about a voltage levelV_(L) is shown by range Δ 4730. In an embodiment, hysteresis is employedto help achieve a PLL lock condition 4732 corresponding to a frequencyF₁ at control voltage level V_(L) corresponding to a tuning capacitancevalue C₂.

In an alternate embodiment the utilization of a hysteresischaracteristic built into a comparator circuit aids in maintaining phaselock. If a single fixed threshold V₁ is used, and a lock is attemptedduring a temperature change, it is possible that a phase lock conditionfor the loop would not be obtainable. For example, if lock at 900 MHz isbeing attempted, the circuit hunts along one of the families of curvesdefined by various numbers of capacitors being switched into thecircuit. The intersection of the vertical line extending through V_(L)and the horizontal line extending from 900 MHz defines the point atwhich lock is desired. Using a well defined V₁, has the problem thatcontrol voltage may be swept along a capacitance curve and past the lockpoint without producing a lock. The process would then switchcapacitance in or out of the circuit causing a jump to a new curve ofthe family tending to pass the lock point without locking the PLL.Hysteresis tends to force the process to hunt along the presentlyselected curve for a slightly longer time to ensure that the PLL lockswhile on the correct capacitance curve.

FIG. 47d is a graph of a family of frequency versus control voltage forvarious capacitor values that illustrates the use of dual comparatorwindows to aid in achieving a frequency lock condition. The graphillustrates the sliding window of valid lock ranges provided by thedesign. A valid lock range for a low V_(GT) and a high V_(GT) are shown.The voltage range of the window is constant. However, the starting andending values of the window vary.

Once the fine or narrow band PLL has been tuned such that it has beenlocked, its frequency may be used in conjunction with the frequencygenerated by the coarse PLL to provide channel tuning as previouslydescribed for the coarse/fine PLL tuning of FIGS. 21 and 22.

Receiver

FIG. 48 is a block diagram of a first exemplary embodiment of areceiver. FIGS. 48, 51, 52, 53 and 54 are embodiments of receivers thatutilize band pass filters and image reject mixers to achieve imagerejection that tend to reduce the distortion previously described. Theembodiments advantageously convert an input signal (1906 of FIGS. 19,48, 51, 52, 53 and 54) to a final IF frequency (1914 of FIGS. 1948, 51,52, 53 and 54) by processing the input signal substantially as shown inFIG. 19. Image rejection is measured relative to the signal strength ofthe desired signal. The strength of the unwanted image frequency ismeasured in units of decibels below the desired carrier (dB_(c)). In theexemplary embodiments of the invention an image frequency rejection of60 to 65 dB_(c) is required. In the embodiments of the invention thisrequirement has been split more or less equally among a series ofcascaded filter banks and mixers following the filters. The filter banks1912,1934 provide 30 to 35 dB_(c) image rejection and complex mixers4802,4806 used provide an additional 30 to 35 dB_(c) of image rejectionyielding an overall image rejection of 60 to 70 dB_(c) for thecombination. The use of complex mixing, advantageously allows therejection requirements on the filters to be relaxed. First, a channel ofan input spectrum is centered about a first IF frequency.

FIG. 49 is an exemplary illustration of the frequency planning utilizedin the embodiments of the invention for the reception of CATV signals.The frequency spectrum at the top of the FIG. 4902 illustrates exemplaryreceived RF signals ranging from 50 to 860 MHz 4904. The received RFsignals are applied to a band pass filter 4921 to eliminate out of banddistortion products Image1 4906. The frequency plan advantageouslyutilizes a trade off between image rejection achievable by filters andmixers at different frequencies. The processing of the first IF and thesecond IF have many features in common and will be discussed together inthe following paragraphs.

For example, the second mixer 4802 and second bank of IF filters 4834 ofFIG. 48 achieve 35 dB and 35 dB of image rejection, respectively. Thethird mixer 4806 and the third IF filter bank 1936 of FIG. 48 achieve 25dB and 40 dB of image rejection respectively. The last distributionreflects the fact that at the lower third IF frequency the Q of thefilters tend to be lower, and the image rejection of the mixers tend tobe improved at lower frequencies.

For example, returning to FIG. 48, a signal 1906 in the 50 to 860 MHzrange is up converted by mixer 1916 and LO2 1908 to 1,200 MHz IF-1 1918.The presence of LO-2 1904 at 925 MHz that is required to mix the signalIF-1 1918 down to the 275 MHz IF-2 1922 has an image frequency Image2(4908 as shown in FIG. 49) at 650 MHz. The filter Q of the 1,200 MHzcenter frequency LC filter 1912 causes Image2 to undergo 35 dB ofrejection thus, attenuating it. To achieve 70 dB of image rejectionanother 35 dB of rejection must be provided by the second mixer (4702 ofFIG. 48) that converts the signal from 1,200 MHz to 275 MHz.

Continuing with FIG. 48, the same structure as described in thepreceding paragraph is again encountered, but at a lower frequency forthe second IF 4914. Image rejection of the 275 MHz filter (1934 of FIG.48) is less due to its lower Q and the fact that the image frequencyImage3 4912 is spaced only 88 MHz 4910 from the signal IF-2 4914. In theprevious first IF stage the image frequency Image2 4908 was spaced 550MHz 4918 from the signal IF-1 4916, providing better image attenuationby filter stop bands. In this situation 25 dB of selectivity can beachieved in the filter, requiring 40 dB of rejection in the mixer toachieve at least 65 dB of attenuation of Image3.

Phase matching at lower frequencies is more accurate allowing betterimage rejection to be obtained from the third mixer. The method oftrading off filter selectivity against mixer image rejection atdifferent frequencies advantageously allows a receiver to successfulintegrate the filters on chip with the desired image frequencyrejection. This process is described in detail in the followingparagraphs.

Returning to FIG. 48, it is desired to up convert a channel received inthis band of signals 1906 to a channel centered at an intermediatefrequency of 1,200 MHz 1918. A local oscillator 1908 producesfrequencies from 1,250 MHz to 2060 MHz. For example, a channel centeredat 50 MHz is mixed with the local oscillator set at 1,250 MHz to producefirst IF frequency components 1918 at 1,200 MHz and 1,300 MHz. Only oneof the two frequency components containing identical informationproduced by the mixing process is needed; the low side 1,200 MHzcomponent is kept. Filtering 1912 tends to remove the unneeded high sidecomponent and other desired signals.

Choosing the first IF 1918 to be centered at 1,200 MHz makes the firstIF susceptible to interference from a range of first image frequenciesfrom 2,450 MHz to 3,260 MHz (4906 as shown in FIG. 49), depending uponthe channel tuned. The lower image frequency of 2,450 MHz results fromthe first IF of 1,200 MHz being added to the lowest first LO present at1,250 MHz to yield 2,450 MHz. The highest image frequency results fromthe first IF of 1,200 MHz being added to the highest first LO of 2,060MHz to yield 3,260 MHz as the highest first image. Choosing the first IF1918 at 1,200 MHz yields image frequencies (4906 of FIG. 49) that arewell out of the band of the receiver. The result tends to placeundesired frequencies far down on the filter skirts of filters presentin the receiver, attenuating them.

After a channel is up conversion to a first IF 1918 of 1,200 MHz, it isnext filtered by a bank of 3 LC band pass filters 1912 each having itsresponse centered at 1,200 MHz in the embodiment. These filters inconjunction with the second mixer 4802 provide 70 dB of image frequencyrejection (4908 of FIG. 49). Filters are advantageously integrated ontothe CMOS substrate. An LC filter comprises inductors (or coils) andcapacitors. An inductor implemented on a CMOS substrate tends to have alow Q. The low Q has the effect of reducing the selectivity and thus theattenuation of signals out of band.

The attenuation of signals out of band can be increased by cascading oneor more filters. Cascading filters with identical response curves hasthe effect of increasing the selectivity, or further attenuating out ofband signals. The embodiments of the invention advantageouslyincorporate active g_(m) stage filters 1912, 1934 to increaseselectivity and provide circuit gain to boost in band signal strength.Three cascaded active LC filters implemented on a CMOS substrate yield asatisfactory in band gain, and provide approximately 35 dB of out ofband image signal rejection in the embodiment described. However, thefilters need not be limited to active LC filters, other characteristicsand passive filters are contemplate equivalents.

The remaining 35 dB of image frequency rejection needed must be achievedin the other circuitry. Hence, differential I/Q mixers 4802,4806 areadvantageously used to achieve this approximate 35 dB of additionalimage rejection required in the first IF.

FIG. 50 is a block diagram illustrating how image frequency cancellationis achieved in an I/Q mixer. An I/Q mixer is a device previouslydeveloped to achieve single side band signal transmission. It is one ofthree known methods for eliminating one of two side bands. This type ofmixer is able to transmit one signal while eliminating or cancelinganother signal. An I/Q mixer advantageously possesses the properties ofimage frequency cancellation in addition to frequency conversion. Forexample, returning to FIG. 48, a second LO 1904 of 925 MHz is used tocreate the down conversion to a second IF 1922 of 275 MHz, whilerejecting image frequencies from the previous frequency conversion byLO1 1908.

The I/Q mixers are implemented in several ways in the invention. Howeverthe overall function is maintained. An interconnection of componentsthat achieves I/Q mixing is illustrated in the exemplary I/Q mixer 4802shown in FIG. 48.

First an input signal 1918 is input to a mixer assembly comprising twoconventional mixers 4828, 4830 of either a differential (as shown) orsingle ended construction.

Local oscillator signals 1904, that need not necessarily be buffered toachieve I/Q mixing, are applied to each mixer. The local oscillatorsignals applied to each mixer are of the same frequency, but 90 degreesout of phase with each other. Thus, one signal is a sine function, andthe other is a cosine at the local oscillator frequency. The 90 degreephase shift can be generated in the I/Q mixer or externally. In thecircuit of FIG. 48 a conventional poly phase circuit 4832 provides thephase shift and splitting of a local oscillator signal generated by PLL24806.

Two IF signals, an I IF signal and a Q IF signal, are output from themixers and fed into another conventional poly phase circuit 4834.Thepoly phase circuit outputs a single differential output IF signal.

Returning to FIG. 50, the I/Q mixer uses two multipliers 5002,5004 andtwo phase shift networks 5006,5008 to implement a trigonometric identitythat results in passing one signal and canceling the other. Thetrigonometric identity utilized is:

cos(2πf _(RF) t)cos(2πf _(LO1) t)±sin(2πf _(RF) t)sin(2πf _(LO1)t)=cos[2π(f _(RF) −f _(LO1))t]  (8)

where

f_(RF) is an input signal 5010

f_(LO1) is the first LO 5012

The signals produced and blocks showing operations to create signaltransformation of these signals to yield the desired final result isshown in FIG. 50. The process makes use of a hardware implementation ofthe trigonometric identities:

sin(u)sin(v)=½[cos(u−v)−cos(u+v)]  (9)

and

cos(u)cos(v)=½[cos(u−v)+cos(u+v)]  (10)

By applying these trigonometric identities to the signals created by thetwo mixers, the product of the sine waves 5014 is:

½[cos(2πf _(LO1) t−2πf _(RF) t)−cos(2πf _(LO1) t+2πf _(RF) t)]  (11)

and the product of the cosines 5016 is:

½[cos(2πf _(LO1) t−2πf _(RF) t)+cos(2πf _(LO1) t+2πf _(RF) t)]  (12)

Thus, two frequencies are created by each multiplication. Two of thefrequencies have the same sign and frequency, so that when they areadded together 5018 the resultant signal is a positive sum 5020. Theother frequency created cancels itself out 5022. The sum frequencycomponent created by the product of the sines is a negative quantity.The same sum frequency component created by the multiplication of thecosines is positive and of equal magnitude. Thus, when these signals areadded together one frequency component, the difference, that is presentin each signal has twice the amplitude of the individual signals and thesecond, sum frequency created is of opposite polarity of the othersignal created and cancels out when the signals are added together.Thus, the difference frequency is passed to the output while the sumfrequency component is canceled.

The implementation of this trigonometric identity by a circuit is veryuseful for canceling image frequencies. As shown in FIG. 4 signal, S andimage signal I are equally spaced by the IF frequency from the localoscillator frequency. The signal frequency would be represented by theterm (2πf_(LO1)t−2πf_(RF)t) and the image frequency would be representedby (2πf_(LO1)t+2πf_(RF)t). In the embodiments of the invention, thephase shifting and summing functions are performed utilizing standardpolyphase or other circuits known in the art.

Mathematically exact cancellation can be achieved. However, real circuitcomponents are not able to achieve exact cancellation of the imagefrequency. Errors in phase occur in the circuitry. A phase error of 3°can yield an image frequency suppression of 31.4 dB_(c) and a phaseerror of 4° can yield an image frequency suppression of 28.9 dB_(c).These phase errors tend to be achievable in an integrated circuit onCMOS. To attempt to achieve the entire 70 dB_(c) of image rejectiontends to be undesirable, thus necessitating the filters. For example, toachieve 59 dB_(c) of image frequency rejection a phase error tending tobe of no more than 0.125° in the mixer would be allowable.

By combining image frequency rejection achievable by an LC filterimplemented in CMOS with an I/Q mixer's image rejection properties,properties that tend to be achievable in a CMOS integrated circuit, arequired image frequency rejection is obtained. Additionally, thefrequency of a first up conversion has been advantageously selected toplace an image frequency of a first LO well down the filter skirts of a1,200 MHz LC filter bank, thus achieving the desired image frequencyrejection.

Returning to FIG. 48, buffer amplifiers 4810 are used to recondition theamplitudes of LO signals 1908, 1904, 1930 that drive the I/Q ports ofmixers 4802,4806. A distance of several millimeters across a chip fromwhere LOs are generated 4504, 4506, 4508, 4502 to where it is applied atthe mixers 1916, 4802, 4806 tends to require reconditioning of theslopes of the local oscillator signals. Buffering also tends to preventloading of the PLLs 4504,4806.

Eliminating any preselection filtering requiring tunable band passfilters is desirable. To do this image frequency response and localoscillator (LO) signals are set to fall outside of a received signalsbandwidth. The first signal conversion tends to eliminate anyrequirements for channel selectivity filtering in the receiver frontend. Because of the integrated circuit approach to this design it isdesirable to locate an LO outside of the signal bandwidth to reducedistortion created by the interaction of the received signals and thefirst local oscillator signals.

An approximately 35 dB of out-of-band channel rejection in the first IFstage's filter 1912 is insufficient. The additional 35 dB of selectivityprovided by a mixer 4802 increases selectivity. However, it is desirableto mix down a received signal as quickly as possible. This is desirablebecause at lower frequencies filters tend to have better selectivitythan at the higher IF frequencies. By converting a received signal to aslow a frequency as possible as quickly as possible better filteringtends to be obtained. Two frequency down conversions are next performed.

Filters are available that will achieve a better rejection than an LCfilter at a given frequency, for example a SAW filter. While betterfiltering of the intermediate frequencies could be obtained with afilter such as a SAW filter at a higher frequency, a fully integratedreceiver would not be achievable. A SAW filter is a piezoelectric devicethat converts an electrical signal to a mechanical vibration signal andthen back to an electrical signal. Filtering is achieved through theinteraction of signal transducers in the conversion process. A filter ofthis type is typically constructed on a zinc oxide (ZnO₂), a materialthat is incompatible with integration on a CMOS circuit utilizing asilicon (Si) substrate. However in alternative embodiments of theinvention, SAW or other filter types known in the art including externalLC filters are contemplate embodiments. In particular, a hybridconstruction utilizing receiver integrated circuit bonded to a hybridsubstrate and filters disposed on the substrate is contemplated.

Returning to the frequency plan of FIG. 49, there is an image response(Image2) 4908 associated with the second local oscillator signal (LO₂)4920. Returning to the embodiment of FIG. 48, this Image2 signal occursat f_(LO2)−f_(IF2)=925 MHz−275 MHz, which is 650 MHz. If there is asignal of 650 MHz at the receiver's input 4808 it is possible that a 650MHz signal will be mixed down to the second IF frequency (IF₂) (1922 ofFIG. 48) causing interference with the desired received signal which isnow located at the second IF frequency. To reduce interference from thissignal the receiver has been designed to produce greater than 65 dB ofrejection of Image2 by the mechanism previously described for the 1,200MHz LC filter bank 1912 of FIG. 48.

Returning to FIG. 48, the third IF is next generated. The third LO 1930is created by direct synthesis. The divide by 4 block 4802 creates a 231MHz third LO (LO₃) consisting of I and Q signals required to mix the 275MHz second IF 1922 down to the third and final IF frequency of 44 MHz1926. A second down conversion to the 275 MHz third IF is used in thedesign. If a 1,200 MHz first IF signal were down converted directly to44 MHz a local oscillator signal of 1156 MHz (1,200 MHz−44 MHz) would berequired. A resulting image frequency for this local oscillator would beat 1,112 MHz (1,200 MHz−88 MHz). A 1,112 MHz image would fall within theband of the 1,200 MHz LC filter. Thus, there would be no rejection ofthis image frequency from the first IF's filter since it falls in thepass hand. Therefore, the intermediate frequency conversion to a secondIF of 275 MHz is used to reduce the effects of the problem.

The 231 MHz third LO 1936 falls close to the center of the receivedsignal band width 1906. With the three frequency conversions of thedesign the third LO necessarily falls within the received signal band.This is undesirable from a design standpoint. This is because anyspurious responses created by a third local oscillator signal fallwithin the received signal bandwidth. The present embodiment of thisinvention advantageously minimizes these undesirable effects.

In generating the third LO signal of 231 MHz, typically a phase lockloop containing a voltage controlled oscillator would be used. However,these frequency components tend to be primary generators of spuriousproducts that tend to be problematic. The present embodiments of theinvention advantageously avoids the use of a PLL and the attendant VCOin producing the third LO signal 1930 at 231 MHz. A divide by 4 circuit4802 utilizes two flip-flops that create the I and Q third LO signals1930 from the 925 MHz second LO 1904. This simple direct synthesis ofthe third LO tends to produce a clean signal. The reduced generation ofdistortion within the signal band tends to be important in an integratedcircuit design where all components are in close physical proximity. Ifa PLL were used to generate the 231 MHz signal an external loop filterfor the PLL would be utilized, providing another possible path for noiseinjection. By elegantly generating this third LO, that necessarily fallswithin the received signal bandwidth, noise and interference injectionthrough the substrate into the received signal path tends to beminimized.

LC filter tuning 4812, 4814, 4816 in the embodiment is advantageouslyperformed at startup of the chip. A “1,200 MHz filter tuning” circuit4812 tunes the 1,200 MHz low pass filters 1912; a “275 MHz filtertuning” circuit 4814 tunes the 275 MHz low pass filter 1934; and a“44/36 MHz filter tuning” circuit 4816 alternatively tunes a final LCfilter 1936 to one of two possible third IF frequencies (44 MHz or 36MHz) depending upon the application. Alternatively, in this embodiment,the filtering of the third IF frequencies is done by an external filter4818. This external filter may have a saw device or other type of filterthat provides satisfactory filtering of the third IF frequency.

In an embodiment an intermediate frequency automatic gain controlamplifier (“IF AGC”) 3419 is used to provide a nearly constant IFfrequency signal level to IF signal processing/demodulating circuitry(3416 of FIG. 34).

Often the signal level variations being compensated for by the IF AGCare created by improperly tuned filters. The on chip filter tuningutilizing one or more existing PLL signals tends to reduce signal levelvariations.

As previously described, the filter tuning circuits 4812, 4814, 4816utilize tuning signals based on the PLL2 signal 4806, with the “44/36MHz filter tuning” circuit utilizing the PLL2 frequency divided by four4802. However, the tuning signals selected may vary. Any or all of thePLLs 4804, 4806, 4802 or reference oscillator 4808 may be used togenerate a filter tuning signal. Also a single frequency can be used totune all filters with the appropriate frequency scaling applied. Intuning the LC filters, first the chip is turned on and PLL2 4806 mustlock. PLL2 must first lock at 925 MHz as previously described. A VCO inthe PLL 4806 is centered by adjusting its resonant circuit with tunablecapacitors as previously described.

Once the PLL2 is adjusted to 925 MHz a write signal is sent out toindicate that a stable reference for filter tuning is available. Once astable 925 MHz reference for tuning is available the 1,200 MHz filter,the 275 MHz filter tuning previously described takes place. Once thefilter tuning is finished the filter tuning circuitry sends out a signalover an internal control bus structure, linking the receiver to acontroller indicating that the tuning has finished. The receiver is nowready to select and tune a channel.

Frequency tuning of received channels is accomplished in the embodimentwith a coarse and fine PLL adjustment as previously described. Thetuning is performed in such a way that there is always a third IFpresent at the output during the tuning process. PLL1 4804 is the coarsetuning PLL that tunes in 10 MHz steps. PLL2 4806 is the fine tuning PLLthat tunes in 100 KHz steps. Exemplary tuning steps can be made as smallas 25 KHz. A 100 kHz step is used for QAM modulation, and a 25 KHz stepis used for NTSC modulation.

At the input of the tuner each exemplary channel is separated by 6 MHz.PLL1 jumps in tuning steps of 10 MHz. Therefore, + or −4 MHz is themaximum tuning error. If the filters used had a narrow band passcharacteristic this tuning approach tends to become less desirable. Forexample, if the filter bandwidth was one channel, 6 MHz, wide and thefirst IF could be 1204 MHz or 1196 MHz. Thus, the selected channel wouldnot be tuned. The bandwidth of the cascaded filters in the first IFstrip is approximately 260 MHz. The bandwidth of the filters centered at275 MHz in the second IF strip is approximately 50 MHz. The bandwidthsare set to be several channels wide, a characteristic thatadvantageously takes advantage of the low Q in the LC filters built onthe chip. The two PLLs guarantee that a third IF output is alwaysobtained. The first PLL that tunes coarsely must tune from 1,250 to2,060 MHz, a wide bandwidth. PLL2, the fine tuning PLL, must tune from +to −4 MHz, which tends to be easier to implement.

FIG. 51 shows a second exemplary embodiment of the invention. Thisembodiment is similar to the embodiment of FIG. 48, however iteliminates the first IR reject mixer (4802 of FIG. 48). Theapproximately 35 dB of image rejection that has been eliminated due tothe removal of the IR reject mixer is made up by increased filterrejection provided by a 1,200 MHz LC filter bank 5101. The IR rejectmixer is replaced with a conventional differential mixer 5104. The IOrequired is a single differential LO signal 5106 rather than thedifferential I and Q signals previously described. Better filters areused or alternatively an additional series of three 1,200 MHz LC filters1912 for a total of six cascaded filters 5101 to provide sufficientimage rejection are provided. This design provides the advantage ofbeing simpler to implement on an integrated circuit.

If a higher Q or better filter selectivity is realized on the integratedcircuit 65 dB of image frequency rejection at 650 MHz is required. In analternate embodiment of the invention the third down conversion can beaccomplished in a similar manner by eliminating the third I/Q mixer 4806and increasing the selectivity of the 275 MHz filter bank 5102. Themixer 4806 is replaced with a conventional mixer requiring only a singledifferential third LO.

FIG. 52 shows a third alternate embodiment of the invention that tendsto provide continuous tuning of the filter over temperature, and tendsto more accurately keeps the response curve of the filter centered onthe desired frequency. This embodiment of the invention preserves theseparation of I 5202 and Q 5204 signals through the second IF stage5206. In the third frequency conversion stage 5208 the I and Q signalsare transformed into I′, {overscore (I)}, Q, and {overscore (Q)}signals. This alternate embodiment of the invention relies on a“three-stage poly phase” 5210 to provide image cancellation. Theadvantage of using a gyrator in place of dual LC filter bank 5212 isthat a close relationship between I and Q tends to be maintainedthroughout the circuit. The phase relationship at the output of thegyrator filter tends to be very close to 90°. If an LC filter isutilized there is no cross-coupling to maintain the phase relationshipas in the gyrator. In the LC filter configuration complete reliance uponphase and amplitude matching is relied upon to maintain the I and Qsignal integrity. The gyrator circuit has the additional advantage oftending to improve the phase relationship of signals initially presentedto it that are not exactly in quadrature phase. For example, an I signalthat is initially presented to the gyrator that is 80° out of phase withits Q component has the phase relation continuously improved throughoutthe gyrator such that when the signals exit the gyrator quadrature phaseof 90° tends to be established between the I and Q signals, such as in apolyphase circuit element. This present embodiment of the inventionprovides the additional benefit of being easily integrated onto a CMOSsubstrate since the gyrator eliminates the inductors that an LC filterwould require. Filter timing and frequency generation utilize themethods previously described.

FIG. 53 is a block diagram of an exemplary CATV tuner that incorporatesan embodiment of the present invention. The exemplary embodiments of thereceiver are for terrestrial and cable television reception of signalsfrom 50 to 860 MHz. Television signals in this exemplary band arefrequency QAM or NTSC modulated signals. A receiver as describedperforms equally well in receiving digital or analog signals. However,it is to be understood that the receiver architecture disclosed willfunction equally well regardless of the frequencies used, the type oftransmission, or the type of signal being transmitted. With regard tosignal levels input to the receiver, the dynamic range of the devicesused in the receiver may be adjusted accordingly. Thus, in a wide-bandreceiver distortion products are particularly problematic. The receiverdisclosed in the exemplary embodiments of the present invention tends toadvantageously reduces interference problems created by this type ofdistortion.

In the exemplary embodiments of the invention signals input to thereceiver may range from +10 to +15 dB_(m). Where, zero dBm=10 log(1 mV/1mV). It should be noted that in the case of a cable transmitting the RFsignals, that an attenuation envelope impressed on the signals will havea downward or negative slope. This downward or negative slope is aresult of a low pass filter characteristic of the coaxial cable. Thiseffect may be compensated for by introducing a gain element in thesignal chain that has positive slope, to compensate for the negativeslope resulting from cable transmission.

In a wide band receiver designed to process signals received overmultiple octaves of band width, this transmission characteristic canpresent a problem. For example, in the cable television band going from50 to 860 MHz it is possible for distortion products created by thelower frequency signals in this band width to fall upon one of thehigher tuned frequencies, for example 860 MHz. In a multi octaveband-width receiver harmonic signals are problematic since they alsofall within the receiver band-width, and cannot be low pass filteredout. If a channel at one of the higher frequencies is the desired signalthat the receiver is tuned to, the low pass filter characteristic of thecable, or transmission medium, reduces the strength of this desiredtuned signal relative to the lower frequency untuned signals. Because ofthe relatively greater strength of the lower frequency signal, thestrength of the distortion products generated by them, are comparable instrength to the desired tuned signal. Thus, these distortion productscan cause a great deal of interference with the desired received signalwhen one of their harmonics coincidentally occurs at the same frequencyas the tuned signal.

The frequency plan of this tuner allows it to be implemented in a singleCMOS integrated circuit 4822 and functions as previously described inFIG. 48. This exemplary single up-conversion dual down conversion CATVtuner utilizes two PLLs that run off of a common 10 MHz crystaloscillator 5302. From the 10 MHz crystal oscillator references the PLLsgenerate two local oscillator signals that are used to mix down areceived radio frequency to an intermediate frequency. This integratedCATV tuner advantageously uses differential signals throughout itsarchitecture to achieve superior noise rejection and reduced phasenoise. The receiver of the present invention advantageously provideschannel selectivity and image rejection on the chip to minimize thenoise injected into the received signal path. The differentialconfiguration also tends to suppress noise generated on the CMOSsubstrate as well as external noise that is radiated into thedifferential leads of the 10 MHz crystal that connect it to thesubstrate. In this embodiment, an external front end as previouslydescribed is supplied on a separate chip 5304 and an external filter5306 is utilized.

The details of integrated tuners are disclosed in more detail in U.S.patent application Ser. No. 09/439,101 filed Nov. 12, 1999 (B600:33756)entitled “Fully Integrated Tuner Architecture” by Pieter Vorenkamp,Klaas Bult, Frank Carr, Christopher M. Ward, Ralph Duncan, Tom W. Kwan,James Y. C. Chang and Haideh Khorramabadi; based on U.S. ProvisionalApplication No. 60/108,459 filed Nov. 12, 1998 (B600:33586), the subjectmatter of which is incorporated in this application in its entirety byreference.

Telephony Over Cable Embodiment

FIG. 54 is a block diagram of a low power embodiment of the receiverthat has been configured to receive cable telephony signals. Theseservices among other cable services offered make use of RF receivers. Acable telephone receiver converts an RF signals present on the cable toa baseband signal suitable for processing to an audio, or other type ofsignal routed to a telephone system and a subscriber via two waytransmission. When such services are widely offered, and are packagedinto a common device, per unit cost and power dissipation tend to becomeconcerns. It is desirable to provide a low cost and power efficientreceiver.

Receivers integrated onto a single chip that incorporates filters on thechip reduce cost. However, placing filters onto a an integrated circuitresults in a high power consumption by the chip. On chip filters requiretuning circuitry that tends to consume significant amounts of power.Removal of this circuitry allows reduction of power levels to below 2Watts per receiver. Each time that a signal is routed off of anintegrated circuit the chances of increasing system noise are increaseddue to the susceptibility of the external connections to the pick up ofnoise. Careful signal routing and the proper frequency planning of thepresent embodiment are calculated to reduce these undesired effects.

First, an input signal is passed through an RF front end chip 5304 aspreviously described. The first frequency up conversion to the first IF5402 is performed on the integrated receiver chip. After passing a50-860 MHz signal through a receiver front end 5304 that provides adifferential output to the receiver chip 5404 the signal is downconverted to 1,220 MHz 5402. The 1,270 to 2,080 MHz LO 5406 is generatedon chip by a first PLL circuit, PLL1 5408. The 1220 MHz differentialsignal is passed through buffer amplifiers 5410 and is applied to an offchip differential signal filter 5412, with a center frequency at 1,220MHz having a characteristic impedance of 200 Ohms. The differentialsignal tends to provide the necessary noise rejection when routing thesignal off and subsequently back onto the chip. Next the signal isrouted back on to the integrated circuit 5404 where it is again passedthrough a send buffer amplifier 5414.

The second frequency down conversion to the second IF 5416 is performedon the integrated receiver chip. An 1,176 MHz differential I and Q LO5418 is generated on the integrated circuit by a second PLL, PLL2 5420and polyphase 5422. The resulting second IF frequency 5616 is 44 MHz.The mixer used to generate the second IF is an I/Q type mixer 5424 thatsubsequently passes the signal through a polyphase circuit 5426. Thesecond IF is then passed through a third buffer amplifier 5428. Thesignal is next routed off chip to a differential filter centered at 44MHz 5430. After filtering the signal is returned to the integratedcircuit where it undergoes amplification by a variable gain amplifier5432.

Variable gain amplifier (“VGA”) 5432 utilizes cross coupled differentialpairs as described in FIG. 74. The improved dynamic range of the VGAcompensates for increased variations in signal amplitude caused byirregularities in the external differential filter 5430. By operatingsatisfactorily over a wide dynamic range of input signal levels thefilter requirements may be relaxed, allowing for a more economicalreceiver to be constructed.

The details of a low power receiver design are disclosed in more detailin U.S. patent application Ser. No. 09/439,102 filed Nov. 12, 1999(B600:36232) entitled “System and Method for Providing a Low PowerReceiver Design” by Frank Carr and Pieter Vorenkamp; based on U.S.Provisional Application No. 60/159,726 filed Oct. 15, 1999 (B600:34672),the subject of which is incorporated in this application in its entiretyby reference.

Electronic Circuits Incorporating Embodiments of the Receiver

FIG. 55 shows a set top box 5502 used in receiving cable television(CATV) signals. These boxes typically incorporate a receiver 5504 and adescrambling unit 5506 to allow the subscriber to receive premiumprogramming. Additionally, on a pay for view basis subscribers can orderprogramming through their set top boxes. This function additionallyrequires modulation circuitry and a radio frequency transmitter totransmit the signal over the CATV network 5508.

Set top boxes can, depending on the nature of the network, provide otherservices as well. These devices include, IP telephones, digital set-topcards that fit into PCs, modems that hook up to PCs, Internet TVs, andvideo conferencing systems.

The set-top box is the device that interfaces subscribers with thenetwork and lets them execute the applications that reside on thenetwork. Other devices in the home that may eventually connect with thenetwork include IP telephones, digital set-top cards that fit into PCs,modems that hook up to PCs, Internet TVs, and video conferencingsystems.

To satisfactorily provide digital services requiring high bandwidth, settop boxes must provide a easy to use interface between the user and CATVprovider. Memory 5510 and graphics driven by a CPU 5512 tend to make theapplication as appealing as possible to a user when interfaced with aset top box 5514.

Also the set-top can receive data in Internet Protocol format and has anIP address assigned to it. Also, satisfactory methods of handlingreverse path communications are required to provide interactive digitalservices. All of these services utilize an operating system resident inthe set top box 5502 for providing a user interface and communicatingwith the head end 5514 where the services are provided.

To receive services, and transmit requests for service, bidirectionallyacross a CATV network the data signal must be modulated on a RF carriersignal. The set top box is a convenient place to modulate the carrierfor transmission, or to convert the modulated carrier to a base bandsignal for use at the user's location.

This is accomplished with a radio frequency (RF) transmitter andreceiver, commonly referred to in combination as a transceiver 5508. Abidirectional signal from a cable head end 5514 is transmitted over acable network that comprises cable and wireless data transmission. Atthe subscriber's location a signal 3406 is received an input to thesubscriber's set top box 5502. The signal 3406 is input to a set top boxtransceiver 5504. The set top box transceiver 5504 comprises one or morereceiver and transmitter circuits. The receiver circuits utilized areconstructed according to an embodiment of the invention. From the settop box transceiver, received data is passed to a decryption box 5506.If the television signal has been encrypted, this box performs anecessary descrambling operation on the signal. After being passedthrough the decryption box, the signal next is presented to a set topbox decoder 3416 where the signal is demodulated into audio and videooutputs 3414. The set top box incorporates a CPU 5512 with graphicscapabilities and a memory 5510 to provide an interface and control theset top box through a data transfer structure 5514. An optional inputoutput capability 5516 is provided for a direct user interface with theset top box. To transmit instructions from the user to the head end,information is transmitted over data transfer structure 5514 into thetransceiver module to the internal transmitter via the cable TV networkto the head end.

FIG. 56 is an illustration of the integrated television receiver 5602.This television could be one that processes digital or analog broadcastsignals 5604. An exemplary integrated switchless attenuator and lownoise amplifier 3408 is the first stage in a receiver contained in atelevision set. The integrated switchless attenuator and low noiseamplifier is used as a “front end” of the receiver to adjust theamplitude of the incoming signal. Incoming television signals whetherreceived from a cable or antenna vary widely in strength, from receivedchannel to channel. Differences in signal strength are due to losses inthe transmission path, distance from the transmitter, or head end,obstructions in the signal path, among others.

The front end adjusts the received signal level to an optimum value. Asignal that is too strong produces distortion in the subsequentcircuitry by over driving it into a non linear operating region. Asignal that is too week will be lost in the noise floor when subsequenthigh noise figure circuitry is used in an attempt to boost the signalstrength. When used in conjunction with “automatic level control” (5604)circuitry the integrated switchless attenuator and low noise amplifierresponds to a generated feed back signal input to its control voltageterminal to adjust the input signal level to provide optimumperformance.

After passing through the front end 3408, the RF signals 5604 are inputto tuner 5620. This tuner circuit is as described in the previousembodiments where a single channel is selected from a variety ofchannels presented in the input signal 5604. An automatic fine tuningcircuit (“AFT”) 4622 is provided to adjust the level of the final IFsignal 5624 being output to the television signal processing circuitry5610. The signal processing circuitry splits the audio signal 5602 offof the final IF signal 5624 and outputs it to an audio output circuitsuch as an amplifier and then to a speaker 5618. The video signal splitfrom IF signal 5624 is delivered via video signal 5606 to videoprocessing circuitry 5612. Here the analog or digital video signal isprocessed for application as control signals to the circuitry 5614 thatcontrols the generation of an image on a display device 5626. Such areceiver would typically be contained in a television set, a set topbox, a VCR, a cable modem, or any kind of tuner arrangement.

FIG. 57 is a block diagram of a VCR that incorporates an integratedreceiver embodiment 5702 in its circuitry. VCRs are manufactured withconnections that allow reception and conversion of a televisionbroadcast signal 5704 to a video signal 5706. The broadcast signals aredemodulated 5708 in the VCR and recorded 5710 on a recording media suchas a tape, or output as a video signal directly. VCRs are a commodityitem. Cost pressures require economical high performance circuitry forthese units to provide additional more features as the prices decline inthe marketplace.

FIG. 58 shows a block diagram of a typical cable modem. A “Cable Modem”is a device that allows high speed data connection (such as to theInternet) via a cable TV (CATV) network 5812. A cable modem commonly hastwo connections, one to the cable TV wall outlet 5802 and the other to acomputer 5804.

There are several methods for connecting cable modems to computers,Ethernet 10BaseT is an example. The coax cable 5808 connects to thecable modem 5806, which in turn connects to an Ethernet card 5814 in aPC. The function of the cable modem is to connect broadband (i.e., thecable television network) to Ethernet. Once the Ethernet card has beeninstalled, the TCP/IP software is typically used to manage theconnection.

On-line access through cable modems allows PC users to downloadinformation at a speeds approximately 1,000 times faster than withtelephone modems. Cable modem speeds range from 500 Kbps to 10 Mbps.Typically, a cable modem sends and receives data in two slightlydifferent, or asynchronous fashions.

Data transmitted downstream, to the user, is digital data modulated ontoa typical 6 MHz channel on a television carrier, between 42 MHz and 750MHz. Two possible modulation techniques are QPSK (allowing datatransmission of up to 10 Mbps) and QAM64 (allowing data transmission ofup to 36 Mbps). The data signal can be placed in a 6 MHz channeladjacent to an existing TV signals without disturbing the cabletelevision video signals.

The upstream channel to the ISP provider is transmitted at a ratebetween 5 and 40 MHz. This transmission path tends to inject more noisethan the downstream path. Due to this problem, QPSK or a similarmodulation scheme in the upstream direction is desirable due to noiseimmunity above that available in other modulation schemes. However, QPSKis “slower” than QAM.

Cable modems can be configured to incorporate many desirable features inaddition to high speed. Cable modems can be configured to include, butare not limited to, a modem, a tuner 5816, an encryption/decryptiondevice, a bridge, a router, a NIC card, SNMP agent, and an Ethernet hub.

To transmit and receive the data onto the cable television channel itmust be modulated and demodulated respectively. This is accomplishedwith a radio frequency (RF) transmitter and receiver, commonly referredto in combination as a transceiver 5818. The receiver's front end 5820is advantageously provided as previously described.

ESD Protection

FIG. 59 is an illustration of a typical integrated circuit die layout.An IC die 5900 is typically laid out with a series of pads 5904 at theedge of the die. This peripheral area of the die is referred to as thepad ring 5906. Typically at the center of the die a core 5902 islocated. The core contains the circuit functions being performed on theintegrated circuit die 5900. An integrated circuit die is typicallyplaced inside of an IC package or “header”. The IC package provides amechanically sturdy package to protect the die 5900 and interfacereliably with external circuitry. The pads 5904 in the pad ring 5906 aretypically wire bonded to pins fixed in the header. Arranging pads 5904in a peripheral pad ring 5906 allows for ease in an automated wirebonding from header pins to the pads of the die 5900.

Thus, on an IC die 5900, typically configured as shown in FIG. 59, thepads 5904 located in the pad ring 5906 are an intermediate connectionbetween the circuit core 5902 and outside connections on the IC package.

The pad ring of an integrated circuit die typically provides aconvenient place to provide electrostatic discharge (“ESD”) protectioncircuitry. ESD discharge occurs when static build-up of electricalcharge occurs. A static charge build-up typically comprises a highvoltage until discharged. A static charge built up upon a surface willjump, or arc, to another surface of lower potential once the voltagedifference between the surfaces exceeds a spark gap voltage for adielectric, that separates the two surfaces. Spark gap voltages aretypically rated in volts per inch. This is the voltage required to arcfrom one surface to another, located one inch away from each other witha given material present between the surfaces. For a given separatingmaterial a charge will arc from one surface to the other for a lowervalue of potential if the surfaces are moved closer together. Inintegrated circuits distances between conductors or devices present onan integrated circuit tend to decrease as the degree of miniaturizationincreases. Thus, electrostatic discharge from one surface to anotherwithin an integrated circuit tends to occur at smaller voltages as thestate of the art advances.

ESD is a major source of integrated circuit damage. After a chargebuilds up to a point where it arcs from one surface to another, thearcing causes damage to the integrated circuit. Typical damage comprisesholes punched in a substrate and destruction of transistors in the core5902.

ESD protection is typically provided by a device that provides a lowimpedance discharge path from an IC pin to all other pins includingground when an ESD charge exceeds a predesigned threshold voltage of theprotection device. During normal operation of the circuit the ESD devicedoes not cause a loading at the IC pin. Better ESD protection tends tobe produced when a lower trigger threshold is provided in the ESDprotection circuit. (ESD circuits provide a low impedance discharge pathfrom any pin of an integrated circuit to any other pin once an ESDtriggers a given threshold designed into an ESD circuit). Thus, toprotect integrated circuits from ESD signal isolation from pin to pin isundesirable. To withstand an ESD event, large structures with sufficientspacing tend to provide increased ESD protection.

However, from a signal isolation prospective, it is desirable to have ahigh signal isolation between integrated circuits pins. Isolationbetween pins is particularly desirable in RF integrated circuits. Tofunction properly, circuits tend to require power supply lines, groundlines and signal lines that are isolated. ESD circuitry conflictinglytends to require all pins to be interconnected somehow. Furthermore, RFIC's tend to need small structures in order to enhance bandwidth andreduce noise. This requirement is contradictory to an ESD's circuitsrequirement for structures that handle large currents.

An increasing trend in integrated circuit design is to mix high speedand/or high frequency circuitry with high digital circuits. Digitalcircuits tend to generate high noise levels within an IC. Digitalcircuit noise tends to interfere with other circuit functions present onthe die. The individual circuits present on the die are often designedin blocks that define a given area on the die substrate. These circuitblocks containing sensitive circuitry are shielded as much as possiblefrom the digital circuitry.

A common technique to minimize noise injection is to put differentcircuit blocks on separate power and ground lines. Sensitive circuits inthis arrangement are placed as far as possible from noisy circuitry.While this arrangement tends to improve power supply and groundisolation, ESD discharge problems tend to be aggravated.

During ESD discharge a current flows from one to point to anotherthrough path of least resistance. If a path is not present, orinadequate, parasitic discharge paths tend to form causing damage to theintegrated circuit. Thus, circuitry separated by large distances tominimize cross talk and noise injection tend to be susceptible to damagefrom ESD discharge over parasitic paths.

For example, for a noise sensitive mixed mode IC fabricated by a CMOStechnology, a non-epitaxial process is preferred due to the processesability to provide a higher substrate isolation. However, thenon-epitaxial CMOS process tends to create undesirable ESD dischargepaths due to a triggering of a parasitic bipolar structure inherent withthe process. These discharge paths tend to pass through and damage corecircuitry. Thus, it is desirable to provide a structure that tends tocontrol ESD discharge paths.

From an ESD design standpoint, large ESD structures provide betterprotection than a smaller structure. However, in noise sensitivecircuits, the large ESD structures connected to the circuitry tend toact as noise sources, degrading circuit performance. Thus insertion ofESD structures in noise sensitive circuits must be done with care.

FIG. 60 illustrates an embodiment of the invention that utilizes padring power and ground busses. A pad ring buss utilizes a reference VDD6002 and a reference ground ring 6004 that run through the entire padring of a die along the exterior edge of the die. In an embodiment, thepads 5904 along an edge of the die are arranged in line. In an alternateembodiment, the pads 5904 may be staggered along the edge of the die5900.

The reference VDD rings and reference ground rings serve to connect aseries of localized power domains contained in the core 5902 of the die.Because of the block structure making up individual circuit functionswithin the core comprise localized power domains they connect to aprimary power bus in the pad rings. The pad rings 6002, 6004 may bebroken 6006 to prevent the formation of a current loop causing eddycurrents. The pad rings are connected to individual power domains withinthe circuit through ESD discharge protection structures.

FIG. 61 is an illustration of the connection of a series of powerdomains 6102, 6104, 6106 to a pad ring bus structure 6002, 6004. On die5900 pad rings 6002, 6004 are disposed about the periphery of anintegrated circuit. The pad rings are provided with a gap 6006. The padrings surround an integrated circuit core 5902 that comprises one ormore circuit blocks 6102, 6104, 6106. Within each block a localizedpower and ground bus structure is provided for each block 6110, 6112,6114 respectively. ESD discharge protection devices 6108 are utilized toprevent electrostatic discharge damage.

The localized bus structures 6110, 6112, 6114 are connected through ESDdischarge protection devices to the pad rings at a single point. In thisstructure, no localized power supply or ground line is more than two ESDstructures away in potential drop from any other voltage or groundstructure.

FIG. 62 is an illustration of an embodiment utilizing an ESD ground ring6200. In the embodiment shown a set of localized power and ground buses6110, 6112, 6114 are located in a corresponding circuit function blocks6102, 6104, 6106. It is understood that the localized power and groundbusses may contain multiple power and ground lines, and that forsimplicity in explanation a single power supply line and ground linewill be discussed. It is also understood that any number of circuitfunction blocks may be utilized in the circuit to provide the desiredprotection. The circuit function blocks are protected from ESD byutilizing the ESD ground ring 6200 coupled to a series of ESD protectiondevices 6204, 6108.

Each of the localized power and ground busses being protected isconfigured as in circuit function block 6102. The interconnections incircuit block 6102 will be discussed as a representative example of allconnections. A discharge path for power supply lines is through the ESDprotection device 6108 coupled between a local power line VDD1 and alocal ground line GND1. The ESD ground ring and ESD protection devicesprovide isolation between the voltage buss and ground within the circuitblocks 6102, 6104, 6106. The structure also provides an ESD dischargepath between any voltage bus line contained in another circuit functionblock and ground.

Local grounds 6110, 6112, 6114 are coupled through an ESD clampstructure 6204 to the ESD ground ring. To prevent eddy currents fromforming, a gap 6006 is cut in the ESD ground ring 6200. A bond pad 6202coupled to the ESD ground 6200 is provided to couple the ESD ground to asystem ground. Coupling an ESD ground to a system ground tends todecrease noise that tends to be coupled through the ESD ground ring intothe circuit core 5902.

In each circuit function block all individual grounds Gnd1 Gnd2 Gnd3 areconnected to the ESD ground ring through a pair of anti-parallel diodes6204. In addition to anti parallel diodes other ESD triggered protectiondevices may be equivalently utilized. Thus, with the connectiondescribed, any ground in any circuit block is only two diode potentialdrops (approximately 0.6 of a volt for a silicon diode) away from anyother ground in any circuit block.

When implemented in a CMOS technology the substrate is conductive. InCMOS technology the ground lines in each block are inherently coupled toeach other through the substrate. By going through the ESD ground ringthe localized grounds tend to be loosely coupled to each other throughthe pair of anti-parallel diodes. Because of loose coupling between thesubstrate and ESD ground ring, noise coupling between the variousgrounds tends to be minimized.

The VDD lines in each block are completely isolated from each other. TheESD clamps 6108 between the VDD and ground lines in the circuit blocktend to provide a complete discharge path for the VDD bus lines. When anESD event occurs the VDD supply lines in a block sees a low impedancepath through two diodes and two ESD clamps to the VDD bus of anothercircuit block.

RF and high speed signals present unique problems to providing ESDprotection. Noise is typically injected in a circuit through thecircuit's power supply and ground leads. Good high impedance RFisolation of noise sources from an RF signal while providing a lowimpedance ESD discharge path is provided by circuitry comprising an ESDpad ring. The embodiments tend to provide isolation of RF signals fromnoise sources by high impedance paths between the noise signal and RFsignal while maintaining a low impedance discharge path from pin to pinof the integrated circuit when presented with an ESD signal. Thus, thedual requirement of an RF signal's need for isolation and an ESDcircuit's needs for all pins to be connected tends to be achieved in theembodiments described above.

Another conflicting requirement is an RF circuit's need to maintainsmall structures that reduce noise coupling and enhance bandwidth byreducing parasitic capacitance verses an ESD circuit's requirements fora large structure that will withstand a large ESD discharge current.

FIG. 63 is an illustration of the effect of parasitic circuit elementson an RF input signal. Parasitic effects tend to be more pronounced in acircuit structure with large physical dimensions such as a bonding pad.In a typical RF integrated circuit a bonding pad tends to havedimensions much greater than the circuit elements present on theintegrated circuit. In addition bonding pads are attached to pins of anintegrated circuit often by wire bonds that increase the parasiticeffects. Parasitic elements tend to produce the affects of a low passfilter 6300. For simplicity the low pass filter is shown as a seriesresistor 6302 with a shunt capacitance 6304. However in an actualcircuit it is understood that this resistance and capacitance comprisesdistributed elements disposed along the length of the bond wire and padstructure.

If an RF signal 6306 having a given bandwidth is presented to such afiltering structure 6300, then the signal emerging at the other end is aband limited or filtered signal 6308. Such a distorted signal isundesirable. In the case of an analog RF input signal information, orthe signal its self may be lost. In the case of a digital signal,limiting the bandwidth of the spectral components that make up the pulsetrain causes distortion in the pulse train at the output. Thecapacitance 6304 tends to be produced predominantly by a bonding padstructure that separates the charge collected on the bonding pad from aground underneath it.

In an ESD protection circuit large bonding pads and large ESD structuresare desirable to shunt large ESD currents to ground without damage tothe circuitry. However, when such a large ESD structure or bonding padis present RF signals tend to be degraded due to the parasitic effects.Large capacitance is desirable from an ESD design standpoint. Largecapacitors tend to slow down a buildup of charge, and thus potentialduring an ESD event.

In addition cross-talk is produced by a signal on one line beingcapacitively coupled to a signal on a second line distance between thelines must be maintained. A reference ring routed about the periphery ofa chip with bonding pads placed on the core side tends to reduce oreliminate the cross-talk that would occur between these conductors ifone were routed on top of the other.

Returning to FIG. 59, in the state of the art power buses are typicallydisposed between the integrated circuit core 5902 and the pad ring 5906,with the bonding pads 5904 disposed about the periphery of the chip5900. In this arrangement a pad to core connection typically crosses thepower buses perpendicularly.

FIG. 64 illustrates a cross-talk coupling mechanism. A bonding pad 5904disposed on the periphery of the die 5900 would require interconnectingtraces 6404 to pass over ESD voltage and ground reference pad rings6002, 6004. Any signal present on the integrated circuit track 6404crossing over the ESD protection rings 6002, 6004 are capacitivelycoupled 6402. Signals on reference rings 6002 and 6004 will tend to becoupled onto trace 6404 and vice versa. Thus, it is desirable to placethe bond pad 5904 within the periphery of the reference rings.

In an embodiment bond pads 5904 are disposed within the pad rings 6002,6004. External connections are achieved with bond wire connections thatcross over the pad rings. The crossover gap of the bond wire is muchlarger than the vertical distance between the circuit track 6404 andeither of the reference rings 6002, 6004.

FIG. 65 is an illustration of an ESD device disposed between aconnection to a bonding pad and power supply traces. In a typical IClayout a bonding pad 5904 is connected 6404 to an integrated circuitcore 5902. Traces 6504 typically cross power supply and ground lines6002 6004. An ESD device 6500 is typically disposed between the tracesand the power supply buses. A parasitic capacitance exists between thetraces 6404 and the power supply connections 6002, 6004. This parasiticcapacitance reduces signal bandwidth and degrades noise performancebecause of the low pass filtering affect. Also, with this arrangement acore circuit 5902 must be distanced from the bonding pad 5904 to allowfor the power supply traces 6002, 6004 to pass between the pad and core.This prevents minimization of the distance between bonding pad andcircuit core. Parasitic capacitance between power supply conductors andtraces connecting the core to the bonding pad are not the only problemencountered with this configuration. In the current state of the art thebonding pads tend to increase parasitic capacitance.

FIG. 66 is an illustration of parasitic capacitance in a typical bondingpad arrangement on an integrated circuit. In a typical integratedcircuit a large bonding pad is disposed on the surface of the integratedcircuit die 5900. To prevent pad peeling and liftoff one or more metallayers 6600 are disposed in a layered structure separated bysemiconductor material or oxide. The two metal layers 6602, 6604 shownare coupled to the upper metal layer 5904 by multiple feed-throughs 6606that provide electrical contact and mechanical stability to theuppermost bond pad 5904. With this structure multiple parasiticcapacitance 6610 due to the layout are present. These parasiticcapacitances will couple to the substrate or any circuit traces disposednearby such as a power and ground bus structure.

FIG. 67 is an illustration of a embodiment of a bonding pad arrangementtending to reduce parasitic capacitances. A pad ring bus comprised oflines 6002, 6004, 6200 is disposed about the periphery of the chip 5900.ESD devices 6702 are disposed to the side of a bonding pad 6704. Withthis arrangement a bonding pad 6704 may be connected 6504 to a circuitblock in the core 5902 with a minimum interconnecting trace length. Thepad to core connection 6504 does not overlap any power ground or ESD busstructure. Thus, cross-talk and noise coupling with these structurestends to be minimized. In addition the metal routing width from core tobonding pad is not restricted due to requirements that would be imposedby an ESD structure as described in FIG. 67. In an alternate embodimentthat provides improved ESD handling capabilities, the ESD structures6702 may be increased in size.

In an alternative embodiment the ESD ground bus 6200 is placed at theperiphery of the die. This bus tends to carry noise that is mostdisruptive to circuit operation. Thus, it is desirable to space this busas far as possible from a pad. In the alternate embodiment the groundbus is disposed between the ESD ground bus and the VDD bus to reducecoupling between the ESD ground bus and the VDD bus line.

FIG. 68 illustrates a cross section of the bonding pad structure of FIG.67. The bond pad 5904 is reduced in size to the smallest dimensionallowable for successful product manufacturing. A second metal layer6802, further reduced in area as compared to the top layer, is utilizedas an anchor to hold the bonding pad above it in place during a bondingprocess. With this arrangement a smaller number of feed-throughconnections 6606 are required. By eliminating multiple metal layersbeneath the top layer 5904 a distance between the lower bond pad 6802and the substrate 5900 is increased. As predicted from the capacitanceformula, when the distance is increased between capacitor plates theparasitic capacitance is decreased. The relationship is as follows:

C=K∈r×(A/d)  (13)

where

C=capacitance

K=dielectric constant

∈r=the relative dielectric constant of the separating material

A=area of the conducting plates

d=distance between the conducting plates

As can also be seen from the equation the reduced area of the bondingpad results in a smaller capacitance. In addition, if the dielectricconstant in the equation is lowered then the capacitance will also belowered.

A diffusion area 6804 is disposed beneath the bonding pads 5904,6802 todecrease the capacitance from bonding pad to substrate. The diffusionarea comprises a salicided diffusion implant 6804 to further reduceparasitic capacitance coupling to the substrate. This diffusion area6804 is coupled to a potential 6806 that tends to reduce a voltagedifference between the diffusion layer 6804 and the bond pad structure5904, 6802.

FIGS. 69a-69 e illustrate various ESD protection schemes utilized in thestate of the art to protect an integrated circuit from ESD discharge dueto charge build up on a die pad. Typically a large ESD structure (orclamping device) attached to an IO pin of a CMOS integrated circuitallows a high ESD discharge current to be shunted to ground through it.However, a large ESD structure on an IO pin causes two problems. Firstdedicating a large area on an integrated circuit die to an ESD structureis undesirable. Die size is directly related to the cost ofmanufacturing making a minimized die size desirable. A second problemwith a large ESD structure is a capacitive loading by the ESD structureon a signal present on the pin. The loading causes a decrease inbandwidth of the input signal, increased power dissipation, andexceeding the allowable specified input capacitance. A compact ESDprotection structure that works in conjunction with over-voltageprotection, has a fast response time, will not be turned on by noisegenerated in normal operation, and provides a layout that may be used bymultiple semiconductor foundries is described in the followingparagraphs.

In the past various structures 6902, 6904, 6906, 6908, 6910 have beencoupled to IC die pads 5904 to shunt away harmful ESD levels. A commonstructure is the ggNMOS ESD structure 6902. A ggNMOS transistor M1 isutilized to shunt an ESD charge to ground. The source of M1 is tied tothe pad, and the drain to ground. Equivalently the drain may be tied toa lower potential source. As ESD charge builds on the pad its voltageincreases to a point where the ggNMOS transistor is triggered to conductthe ESD charge to ground.

Internal capacitance in the ggNMOS transistor feeds a portion of thevoltage established by a static charge to the ggNMOS transistor gate.When the voltage has risen to a sufficient level on the gate thetransistor conducts. When conducting the transistor is in a lowimpedance state and all the static charge on the pad is shunted toground.

Until the gate voltage rises to a level to cause the transistor toconduct it is in an off, or high impedance state. In this state theggNMOS transistor tends to not disturb the signal on the pad.

Gate bias determines the effectiveness of this structure. In normaloperation the gate of the ggNMOS is biased off putting the NMOS in anoff, or high impedance state. Under an ESD discharge condition the gateof the ggNMOS is biased high to turn on a channel under the gate oxide.The ggNMOS relies on the transistor's inherent capacitance from gate todrain (“Cgd”) to pull the gate high when the pad is pulled high when alarge electrostatic charge is present. Triggering is set by a voltagedivider circuit comprising Cgd and resistor R. The electrostatic chargeon the pad 5904 is divided down by the ratio of impedances of thecapacitor Cgd and resistor R.

Coupling through Cgd degrades in a typical cascode over-voltageprotection circuit. The ggNMOS cannot be used alone without a seriescascode transistor 6904 when its voltage from drain to source (“VDS”)exceeds a given electrical overstress limit. The ggNMOS M1 utilizes aseries cascode stage M5, with its gate biased on, as shown at 6904prevents Cgd from being directly coupled to a bonding pad 5904,substantially impairing its effectiveness. To circumvent insufficientcoupling of M1's Cgd to the pad three other device configurations 6906,6908, 6910 are known.

The first device 6906 adds capacitor C1 to the ggNMOS structure of 6902.C1 is coupled from gate to source of M1. C1 increases the couplingeffect produced by the inherent Cgd of the ggNMOS. Unfortunately C1strongly couples the ggNMOS to the pad. Slight perturbations present onthe pad during normal operation are directly coupled to the ggNMOSthrough the strong coupling. Thus, with the added coupling capacitor C1present, typical AC noise present on the pad tends to turn on the ggNMOSduring normal operation.

The next circuit 6908 utilizes the same coupling capacitor C1 asdescribed in 6906. However, this coupling capacitor has one terminaltied to the gate of M1 and the second terminal tied to a power supplyvoltage. During an ESD event the power supply is pulled high by the ESDvoltage present on the pad. When the power supply is pulled high thegate of the ggNMOS M1 follows it to a high state. However with thisarrangement the gate of the ggNMOS is directly coupled to a noisetypically present on a power supply line. Switching noise present on apower supply line tends to cause the ggNMOS M1 to turn on. If a quiet,or filtered, power supply is coupled to capacitor C1 an extra voltagedrop caused by going through ESD protections of the quiet power supplywould be required before the gate bias is pulled high. This causes anundesirably slow response time.

The third method 6910 utilizes a zener diode Z1 connected with thepositive terminal at the gate of M1 and its negative terminal to thesource of M1 to pull the gate of the ggNMOS high under an ESD discharge.When an ESD discharge event occurs the zener diode goes into a voltagebreakdown mode allowing charge to flow to the gate of the ggNMOS M1. Thegate floats high and the ggNMOS turns on shunting the ESD current toground. The drawback of this approach is that zener diodes are notavailable in standard digital CMOS process.

FIG. 70 illustrates an approach to pad protection during ESD event.Electrostatic charge builds up on an integrated circuit pad 5904. Ashunt device 7002 is connected from the pin 5904 to ground. The shuntdevice 7002 is in a high impedance state until sufficient charge buildsup upon the pad 5904 to trigger the shunt device into a low impedancestate. A low impedance state allows all of the charge built up upon thepad to be shunted to ground before damage to circuitry coupled to thepad can occur. The shunt device is triggered by the ESD charge buildingon the pad. A divider circuit comprising a capacitive element 7006 inseries with a resistive element 7004 are coupled between the pad 5904and ground. The junction of the capacitive and resistive element is usedas a trigger to the shunt device 7002. When a preset trigger voltage isreached the shunt device is activated into a low impedance state.

FIG. 71 is a schematic of a circuit immune to noise that uses an ggNMOS′Cgd and a gate boosting structure to trigger ESD protection. In thisconfiguration diode CR1, transistors M2 and M3 are all disposed in ann-well biased at a voltage V to form a gate boosting structure 7102. Thesource and drain of M3 are coupled to the n-well 710. The source oftransistor M2 is tied to a quiet power supply V. Power supply V is usedto provide back gate bias in the N-well. CR1 is made by a P+ diffusioninto the n-well. Typically only one quiet power supply is sufficient tobias the entire chip. This is because CR1 is fabricated with smalldimensions and dissipates little power.

Transistor M3 is a PMOS transistor operating in its linear region toprovide a MOS capacitor inherent to its construction between CR1 and R1.The drain of M2 is coupled to the source of M3. The drain of M3 iscoupled to the negative terminal of CR1. The positive terminal CR1 iscoupled to the pad 5904. The gate of M3 is coupled to a first terminalof resistor R1, and a second terminal of R1 is coupled to ground. Thejunction of the gate of M3 and R1 is tied to the gate of M1 and thenegative terminal of CR1. The drain of M1 is tied to pin 5904 and thesource of M1 is tied to ground. Alternatively the ground connection isnot at zero potential but some lower potential. Resistor R1 isfabricated as an ohmic resistor, or alternatively using other pulldowntechniques known in the art.

In normal operation M2 is turned on. This provides a low impedance pathfrom the n-well back gate 7100 which is the n-well that host 7102 to thequiet power supply V. The channel side, that is formed by the gate andconductive channel formed in the silicon between the source and drain,of the MOS capacitor formed by M3 is thus tied to a low impedancesource. Diode D1 is reverse biased forming a high impedance path betweenM3 and pad 5904. Thus, a strong coupling between the MOS capacitorformed by M3 and the pad is not present. Added input capacitance tendsto be negligible by keeping the dimensions of diode CR1 as small asallowed by a process' constraints.

When electrostatic discharge occurs CR1 becomes forward biased,providing a low impedance path from the pad 5904 to the capacitor formedby M3. In response the capacitor formed by M3 charges up, providing a“boosting” to turn on the gate of M1. By providing boosting to the gateof M1 the drain source channel in M1 is turned on quickly forming a lowimpedance connection from the pad 5904 to ground. The fast response timeis particularly suitable for a machine model (“MM”) and charge devicemodel (“CDM”) ESD discharge modes.

The MOS capacitor formed by M3 significantly increases the capacitancepresent on the gate of M1. This allows R1 to be reduced in size tomaintain the same time constant τ (τ=1/R×C) that would otherwise berequired if M3 were absent. Without the presence of the capacitance ofM3, R1 would be required to be in the range of hundreds of kilo-Ohms.Resistors of this value require a large amount of layout area.

Thus R1 and CR1 do not require significant die area. The fabrication ofM3 utilizes thin oxide to form the MOS capacitor also providing acompact layout of this device. M1 is also reduced in size because of thegate boosting provided. In the configuration described, M1 is biased ata higher gate source voltage allowing a channel to conduct current moreefficiently. Thus, a given ESD current is capable of being conducted toground with a smaller transistor M1. The dimensions of M1 do not need tobe made large in order to provide sufficient Cgd for gate boosting,since boosting is primarily accomplished through the capacitancesupplied by M3.

FIG. 72 is a schematic of an alternative embodiment utilizing the gateboosting structure and a cascode configuration. In an I/O applicationthe gate of the cascode transistor is tied directly to a power supplyconnection.

FIG. 73 is a schematic of an embodiment that does not require a quietpower supply. For a small amplitude signal, as in RF signalapplications, the drain to gate coupling of M1 will not turn on thechannel of M1. Under this condition a quiet power supply is notrequired, allowing M2 of FIG. 71 to be eliminated. In this embodimentthe pad is coupled to a silicon substrate through the N-well capacitanceof diode CR2. The PMOS capacitor M3 of FIG. 71 is replaced by a metalcapacitor that reduces total n-well area coupled through CR2. Theconfiguration further reduces pad capacitance while still allowing gateboosting of shunting transistor M1 during an ESD discharge.

The details of ESD protection are disclosed in more detail in U.S.patent application Ser. No. 09/483,551 filed Jan. 14, 2000 (B600:34208)entitled “System and Method for ESD Protection” by Agnes N. Woo, KennethR. Kindsfater and Fang Lu based on U.S. Provisional Application No.60/116,003 filed Jan. 15, 1999; U.S. Provisional Application No.60/117,322 filed Jan. 26, 1999; and U.S. Provisional Application No.60/122,754 filed Feb. 25, 1999; the subject matters of which areincorporated in this application in their entirety by reference.

IF AGC Amplifier

The VGA and PGA/LNA have characteristics in common that allowinterchangeability in alternative embodiments.

FIG. 74 is a block diagram of a variable gain amplifier (“VGA”) 3403.The VGA produces a signal that is a reproduction of a signal input to itat an amplified level. The amplified level in a VGA is capable of beingvaried. A variable gain is accomplished through the use of one or morecontrol signals applied to the amplifier.

VGAs are frequently used to maintain a constant output signal level.VGAs do this by varying the amplifier gain to compensate for varyinginput levels. In the case of strong or weak signals it is desirable tomaintain a linear gain for input verses output signals with little noiseadded. Maintenance of a linear gain reduces distortion for high levelinput signals. VGAs are often used in IF or RF strips to compensate forprior losses or weak signal reception.

In a linear gain, a 1 dB increase in sinusoidal input signal levelproduces a 1 dB change in the output signal level at that samefrequency. A gain of this nature is termed a “linear response.” If a 1dB change is not produced, this is indicative of an available powerbeing diverted to produce a signal at another frequency of operation. Asignal at a frequency other than desired often interferes with thesignal being amplified and is termed distortion. Thus, the linearity ofan amplifier is a figure of merit, the greater the linearity the betterthe quality of the amplifier. Amplifiers that utilize compensationcircuitry and differential signal transmission tend to have improvedlinearity.

VGA compensation circuitry controls V_(ds). For a large input signal,linearity and low gain is required. With a reduction in V_(ds), goodlinearity and low gain are achieved. If a small signal is input to theamplifier, V_(ds) is increased. The increase in V_(ds) causes one ormore MOSFETs internal to the VGA to be biased in the active region.Active region bias allows for high gain and low noise to be achievedsimultaneously. The VGA utilizes a current steering method of applyingcontrol signals to provide an extended gain range VGA. The control ofV_(ds) allows the production of a linear output when a large signal isapplied to the input.

The VGA has a differential input comprising two signals, +V_(in) and−V_(in) 7408. The VGA has a differential current output comprising twosignals, +I_(out) and −I_(out). In the embodiment shown the differentialcurrent signals are applied to a first and second resistor R1 and R2 toproduce a differential voltage output, +V_(out) and −V_(out) 7410respectively. Equivalently the current outputs may be applied to workagainst any impedance to generate a voltage output.

A set of three control signals 7404 are supplied to the VGA 3403 from alinearization circuit 7402. The linearization circuit 7402 produces thethree control signals 7404 that are derived from a single controlsignal, V_(c) 7406 through compensation circuitry. Control signal V_(c)tends to be proportional to the gain desired in the VGA 3403. The threecontrol signals 7404 control the VGA in a manner such that a desiredgain and a desired linearity tend to be produced by the VGA.

The linearization circuit is stimulated by the control signal V_(c) 7406is supplied by an external DSP chip. The control signal applied to thelinearization circuit 7402 is shaped in a predetermined way. A goal ofshaping the control circuit is to produce the second set of controlsignals 7404 that are applied to the VGA 3403 to produce a desired VGAgain transfer function, measured in decibels, that changes linearly withthe applied control signal V_(c). In the embodiment shown V_(c) is avoltage, however a control circuit may be equivalently supplied. In analternate embodiment the overall transfer function of the VGA isconfigured to yield a linear function of gain as measured with linearunits versus control voltage by appropriately adjusting thelinearization circuit through the application of a log to linearconversion current.

In addition to shaping the gain transfer function, another function ofthe linearization circuit is to control signals that control the VGA toproduce the desired low distortion output. The second set of controlsignals 7404 are shown as a bussed line 7404. The second set of controlsignals comprise a voltage VD1, and a pair of control currents: iSig andiAtten. The second set of control signals 7404 tend to produce a linearchange in gain with variation of the control signal while maintaining anacceptable distortion level in the VGA.

The three control signals are generated by two subcircuits in thelinearization circuit: a current steering circuit and a drain voltagecontrol voltage signal generation circuit. The current steering circuitproduces two signals, iSig and iAtten. The drain voltage control signalvoltage generation circuit produces one signal, VD1.

FIG. 75, is a block diagram of the internal configuration of the VGA andthe linearization circuit. The VGA and linearization circuit toimplement current steering and V_(ds) control of the VGA are describedas a separate function block. However, the functions described may beequivalently merged into the circuit functional blocks of the other.

The VGA 3403 is constructed from two cross coupled differential pairamplifiers 7500 7502. A first differential pair amplifier 7500 includestwo transistors M4 and M10. A second differential pair amplifier 7502includes transistors M13 and M14. The first and second differential pairamplifiers are driven in parallel by a differential input voltage 7408.When referenced to ground, the differential input voltage applied toeach amplifier simultaneously is denoted +V_(in) and −V_(in).

The differential pair amplifiers have differential current outputs +I1,−I1, +I2, −I2, that are combined to produce a differential VGA outputcomprising +I_(out) and −I_(out). The first differential pair amplifier7500 has differential current outputs +I1 and −I1 that are sinusoidaland 180 degrees out of phase from each other. The second differentialpair amplifier 7502 has differential current outputs +I2 and −I2 thatare sinusoidal and 180 degrees out of phase from each other. VGA outputcurrent +I_(out) results from the combination at node 7505 of out ofphase currents −I1 and +I2. VGA output current −I_(out) results from thecombination at node 7507 of out of phase currents +I1 and −I2. Note thatthe currents described above having a minus sign prefix, −I1, −I2, aregenerated in response to input voltage −V_(in), and the currents withplus sign prefixes, +I1, +I2, are generated in response to +V_(in).

A V_(ds) control circuit 7504 within the VGA 3403 supplies a V_(ds),control voltage that is applied to nodes 7505 and 7508. The V_(ds)control circuit receives an input VD1 from a VD1 control signalgeneration circuit 7510 that is a part of the linearization circuit7402. In alternative embodiments the V_(ds) control circuit is mergedinto the VD1 control signal generation circuit 751.

A current steering circuit 7512 in the gain control circuit 7402supplies control signals iSig and iAtten. The signal iSig is a controlinput to the first differential pair amplifier 7500. The signal iAttenis a control input to the second differential pair amplifier 7500.

In the embodiment shown the VGA 3403 is configured to operate at an IFfrequency. However it is understood that the VGA may be configured, byappropriate component selection to function at any desired frequency. Inan IF strip, the addition of a VGA maintains a constant IF output as theinput varies. This is accomplished by adjusting the gain of the VGA. AVGA is useful in any situation where a signal presented to a circuit isof unknown or variable strength.

Functionally the VGA maintains a constant level at its output so thatsubsequent circuitry may be designed that tends to have betterperformance and less noise. In alternate embodiments, the variable gainamplifier may be used at RF or other frequencies to reduce signal levelvariations in a circuit. For example in an embodiment, a VGA 3403 asdescribed may be used in the RF front end 3408 to control the RF signallevel that is applied to a receiver 3402.

The overall gain of the VGA is attributable to the individual gaincontributions of transistors M10 M4, M13 and M14 that produce a currentgain. In an embodiment, the VGA voltage gain is set by providingresistance at the +I_(out) and −I_(out) terminals to establish a voltageoutput, and thus a voltage gain for the amplifier. The exemplaryembodiment includes field effect transistors (“MOSFETs”). Equivalently,other transistor types may be substituted for the MOSFETs utilized inthe exemplary embodiment. A pair of control currents iSig and iAtten anda control voltage VD1 are principally used to provide an extended rangeof available VGA gain and a linear in dB VGA amplifier transfer functionthat provides a desired linearity.

Two methods of gain control are utilized in the exemplary VGA. The firstmethod is V_(ds) control that controls noise and linearity whilereducing VGA gain when large signals are applied, the second is currentsteering that provides an extended range of available VGA gain. The setof three control signals 7404 include iSig, iAtten and VD1.

In the first method of V_(ds) control, gain and linearity in the outputof the VGA tend to be controlled by adjusting each of four transistors'M4, M10, M13, M14 drain source voltages (“V_(ds”)) of the transistors tocontrol a transductance (“g_(m)”) associated with each transistor. If adrain source voltage V_(ds) across a MOSFET device M10, M4, M13, M14 isreduced, a g_(m) transfer characteristic of that transistor, which is afunction of input voltage, becomes flatter. The flatter the g_(m)transfer function the more linearly the transistor tends to operates.The V_(ds) of all four transistors is controlled in order to manipulatean overall g_(m) characteristic for the VGA.

The V_(ds) gain control method tends to reduce VGA output distortion bytending to improve the linearity of the VGA. To improve the linearity,the V_(ds) of the transistors are reduced yielding better linearity inconjunction with a transistor operating point on a flattened g_(m)curve. As an input signal's strength increases, V_(ds) is reducedproviding a linear response VGA. Reducing V_(ds) also tends tocontribute to VGA gain control. For small input signals as V_(ds) isincreased the MOSFETs become biased in the active region where high gainand low noise operation is obtained. The main effect of reducing V_(ds)tends to be control of the linearity of the VGA amplifier.

In the second method, current steering control, currents iSig and iAttentend to set amplifier gain over a large range. An increase in thecontrol current iSig tends to increase gain by causing an increase inoverall amplifier g_(m), while an increase in iAtten tends to decreasegain by causing a subtraction of overall amplifier g_(m). For certaintype and size MOSFETs, the relationship between iSig, iAtten and g_(m)is as shown in equation (14) $\begin{matrix}{g_{m} = {\sqrt{\frac{K}{2}}( {\sqrt{iSig} - \sqrt{iAtten}} )}} & (14)\end{matrix}$

where

iAtten=I_(tot)−Isig

K=a constant of proportionality

For other size/type transistors this relationship may not hold, but theidea is still applicable. The g_(m)s of each transistor M10, M4, M13,M14 is controlled to adjust gain. This is accomplished by subtracting,or adding currents through control lines iSig and iAtten to boost orreduce the VGA g_(m), as required. Control signals iSig and iAttencontrol amplifier gain by adjusting an overall g_(m) of the amplifier. Afixed available control current is available for controlling VGA gainthrough the iSig and iAtten control lines. Gain is controlled byselectively steering the available current into the appropriate controlline. For large VGA signal inputs, the linearity produced in a VGA fromcurrent steering tends to be improved by the addition of the V_(ds)control circuit.

A single stage VGA amplifier with linearization circuitry as describedabove that utilizes current steering and V_(ds) control could yield again control range in excess of 40 dB.

The second method of VGA gain control is V_(ds) control. Linearity inamplifier output tends to be improved by V_(ds) control or “V_(ds)squeezing.” With current steering, no provision is made for improvinglinearity once the input signal becomes large.

Linearity is typically determined by the g_(m) of each of the twodifferential amplifier stages. The first stage comprises M10 and M4. Thesecond stage comprises M13 and M14. The embodiment described tends tohave an increased linearity of 26 dB, corresponding to a factor of 20improvement in linearity over that typically available.

VGA operating conditions determine the distribution the currents iSigand iAtten. When a small signal is applied to the input terminals+V_(in) and −V_(in), it is typically desirable to amplify the signalwith a high gain setting. Transistors M10 and M4 are coupled to thedifferential output so that their g_(m)s tend to contribute to VGAoverall gain. However, transistors M13 and M14 are coupled to the VGAoutput so that their g_(m)s tend to decrease VGA gain through a g_(m)subtraction. Transistors M4 and M10 are controlled by iSig, transistorsM13 and M14 are controlled by iAtten.

For a high gain condition, g_(m) subtraction is undesired.

Thus, for a high gain setting, it is desirable to have most of the gainavailable from devices M10 and M4 contributing to the amplifier'soverall gain. M10 and M4 are set for maximum gain by setting iSig to amaximum current. Correspondingly iAtten is set to a low value ofcurrent. In achieving a maximum gain, a control current is dividedbetween iSig and iAtten such that a maximum current is present in theiSig line.

In the low gain state, the second differential pair transistors M13 andM14 are controlled by iAtten such that they subtract from the gain ofM10 and M4. A large gain present for devices M13 and M14 creates a largegain subtraction in devices M10 and M4 which are controlled by iSig toproduce a minimum gain.

Thus, when the signal input is small, minimum gain on M13 and M14 isdesired and maximum gain on M10 and M4 is desired to produce maximum VGAgain. When the input signal is large, a maximum gain on M13 and M14 isdesired and minimum gain on M10 and M4 is desired to produce minimum VGAgain.

FIG. 76 is a graph of gain versus the control current iSig. Controlcurrent iSig is shown as a fraction of iAtten, with the total currentbeing equal to 1, or 100%. At the far right of the graph, a 0 dBreference is set corresponding to maximum amplifier gain of maximumamplifier g_(m). As iSig is reduced, control current iAtten isincreasing proportionately causing the VGA's overall g_(m) and gain todecrease.

Maximum VGA gain is desirable with a small input signal present at theVGA input. Maximum gain is achieved with a maximum current into the iSigcontrol line and minimum current into the iAtten control line. As thesignal at the VGA input becomes larger, it is desired to decrease theamplifier gain. A reduction in VGA gain is achieved by decreasing thecurrent in the iSig line and increasing the current in the iAttencontrol line. A minimum VGA gain corresponds to maximum current in theiAtten control line and minimum current into the iSig line.

Returning to FIG. 75, the linearization circuit takes the externallysupplied control signal 7406 that is provided as a voltage and convertsit to control signals 7404 that are current and voltage signals. In thecurrent steering circuit 7512 a maximum control signal voltage amplifiedin the embodiment described corresponds to a maximum gain condition withiSig set to a maximum and iAtten being set to a minimum. As the controlvoltage is decreased, iSig decreases and iAtten increases.

The control voltage Vc 7406 is created by digital circuitry that isresponsive to the input level of the amplifier. In the embodimentdescribed the gain control loop is closed in a digital circuitry domainlocated off of chip that produces control signal 7404.

The output of the VGA is sampled to determine if sufficient signalstrength is available for further signal processing. The sample isprocessed by an A to D converter into a digital signal, and the controlvoltage responsive to the level of the VGA output is created.Alternatively, analog methods may be used to sample the output andgenerate control voltage. In an embodiment the VGA is utilized as an IFVGA. In alternate embodiments the VGA is configured for used at otherfrequency bands that require an adjustment in gain.

Stability of the AGC loop is maintained during changes in iSig andiAtten. Stability is achieved in the minimum gain setting by keepingiSig greater than iAtten. In the embodiment described iSig is preventedfrom becoming less than iAtten by the linearization circuit. If iSigbecomes less than iAtten, phase inversion problems tend to occur causinga degradation in VGA performance, disrupting automatic gain control(“AGC”) loop performance in a receiver. This condition is prevented fromhappening by providing appropriate circuitry in the linearizationcircuit.

Also with respect to AGC loop stability, a zero gain setting isundesirable. In the embodiment, the transistors are fabricated withidentical dimensions, and it is possible to set the gain equal to zeroby making the iSig and iAtten currents equal. This is undesirable from acontrol loop stability standpoint. The linearization circuit providesappropriate circuitry preventing this condition from occurring.

Maximum attenuation is determined by how close iSig is allowed toapproach iAtten in value. Thus, the maximum attenuation achieved isdependent upon the stability that is permissible as iSig approachesiAtten.

FIG. 77 is the schematic diagram of an embodiment of the VGA. The VGAhas a control circuit to control the V_(ds) of M10 and M13 at node 7505,and the V_(ds) of M4 and M14 at node 7507.

A control voltage VD1 is generated by the linearization circuit 7510 andapplied to control a differential amplifier U1. The negative input of U1is coupled to node 7505, and the positive input of U1 is coupled to node7507.

A transistor M1 has its source coupled to node 7505, its drain comprisesthe +I_(out) terminal of the VGA. The gate of transistor M1 is coupledto the positive output of U1. A transistor M2 has its source coupled tonode 7507, its drain comprises the −I_(out) terminal of the VGA. Thegate of transistor M2 is coupled to the negative output of A1.

The V_(ds) squeezing is utilized since it tends to improve linearity. Asthe control signal voltage increases, the control voltage VD1 decreasestending to decrease the VGA gain. As previously discussed, iSig isdecreasing and iAtten is increasing to achieve the desired decrease inVGA gain. Concurrently with V_(ds) squeezing, the V_(ds) of all fourtransistors M10, M4, M13, M14 also tends to decrease with increasinginput signal level due to the application of a variable DC voltage atthe transistor source leads. A DC voltage is fixed at nodes 7501 and7503. Thus, the way available to reduce V_(ds) for M10, M4, M13, and M14is to reduce the DC voltage at the +I_(out) and −I_(out) terminals. Avariable voltage source is connected at each node +I_(out) and −I_(out)−7505, 7507.

The sources of M13 and M14 are coupled in common to node 7503 and to thecontrol signal iAtten. Control signal iAtten tends to cause a decreasein amplifier gain, while control signal iSig tends to increase amplifiergain. The sources of M10 and M4 are coupled in common to iSig at node7510. The drains of M10 and M13 are coupled in common to provide anoutput signal +I_(out). The drains of M4 and M14 are coupled in commonto provide an output signal −I_(out). In the exemplary embodiment input−V_(in) is coupled to the gates of M10 and M14. Input +V_(in) is coupledto the gates of M4 and M13. In the exemplary embodiment differentialinputs and outputs are shown in the amplifier. However, it is understoodby those skilled in the art that a single ended configuration isequivalently produced by the use of a device such as a balun.

FIG. 78a illustrates a family of curves showing the relationship of atransistor's drain current (“I_(d)”) to its gate source voltage(“V_(gs)”) measured at each of a series of drain source voltages(“V_(ds)”) from 50 mV to 1 V. From this graph a transconductance, g_(m)is determined. The following relationship defines a g_(m) curve for eachV_(ds) value:

g _(m) =dI _(d) \dV _(gs)  (15)

FIG. 78b is a graph of g_(m) verses V_(gs) as V_(ds) is varied from 50mV to 1 V. To provide improved output linearity performance, it isdesirable to operate a transistor on a curve of g_(m) that has aconstant value and zero slope. As seen in the graph for a V_(ds) ofapproximately 50 mV, the curves of g_(m) verses V_(gs) tend to be flat.As V_(ds) is increased, the curve begins to slope, indicating thepresence of non-linearity in the output signal. As V_(ds) increases thecurve not only begins to slope, but it develops a bow, furthercomplicating the compensation for the non-linearities at the higherlevel of V_(ds). These irregularities in g_(m) tend to be the sources ofnon-linearities in the output of the amplifier. Thus, it is desired toprovide a flat g_(m) response to produce a more linear transfer functionfor the VGA by controlling V_(ds).

FIG. 78c is a graph of the cross-section of FIG. 78b plotting g_(m)verses V_(ds) for various values of V_(gs). As V_(ds) changes fromapproximately 200 mV to 500 mV, g_(m) changes from approximately 5 mS to13 mS. The change in g_(m) from 5 mS to 13 mS by changing V_(ds) may beused to control gain. Thus, as V_(ds) is decreased, the gain isdecreased. Control of V_(ds) provides approximately 9 dB of gain controlrange.

Within the range of V_(ds), graphed between the vertical bars 7801, thevalue for g_(m) remains essentially the same for a range of V_(gs) inputsignal from 1.2 V to 1.4 V. Thus by controlling V_(ds) from 200 mV to600 mV approximately 9 dB of gain control is provided.

When control of V_(ds) is combined with the g_(m) subtraction methodpreviously described, the linear output signal is maintained. Inaddition approximately 8 dB to 9 dB of gain control in addition to thatprovided by g_(m) subtraction contributes to provide overall VGA gaincontrol on the order of 30 dB, in the exemplary embodiment.

Output linearity is often quantitized by measuring an intermodulationproduct produced by two input signals present at differing frequencies(f_(1 and f) ₂ 302 and 304 respectively of FIG. 3). For the VGA a twotone intermodulation (“IM”) product test is utilized, and the distortionas represented by the third order intermodulation product (308 of FIG.3) is measured. Approximately a 26 dB decrease in the third order IMproduct (308 of FIG. 3) tends to be achieved in the exemplaryembodiment.

With the input signal maintained at a constant level, the output signalat +I_(out) and −I_(out) is measured as gain squeezing is performed.Improvement is measured as compared to adjusting gain without utilizinggain squeezing. A reduction in third order intermodulation ofapproximately 25 dB is measured as V_(ds) is squeezed within a range ofapproximately 150 mV to 200 mV. Utilizing a test having two tones at 44MHz and 45 MHz typically produces third order intermodulation productcomponents at 43 MHz and 46 MHz. With this test, 20 dB to 25 dBimprovement in third order intermodulation is observed in the exemplaryembodiment. A typical improvement of 20 dB is realized in the linearityof the output signal.

FIG. 79 is a schematic of a current steering circuit. An externalcontrol signal V_(c) drives a differential pair amplifier 7910 includingMC1, MC2, to ultimately generate iSig and iAtten. The iSig and iAttenare generated through two current mirrors 7904, 7906. The first currentmirror 7904 comprises MC3 and MC6. The second current mirror 7906comprises MC4 and MC5. The circuit maintains a fixed relationshipbetween iSig and iAtten, defined by:

I _(tot) =iSig+iAtten  (16)

To guarantee that phase reversal does not occur, iSig must remaingreater than iAtten at all times. By selecting V_(ref) to be slightlyless than the minimum value of control voltage V_(c) that will bepresent, iSig will remain greater than iAtten.

In an embodiment of current steering circuit 7512, a control voltageV_(c) is applied to a differential pair amplifier 7910. In the exemplaryembodiment, control signal V_(c) ranges from 0.5 V to 2.5 V. The 0.5 Vcorresponds to a minimum gain setting and 2.5 V corresponds to a maximumgain setting. Differential pair amplifier 7910 comprises two transistorsMC1 and MC2. In the exemplary embodiment, field effect transistors areused. Equivalently, other types of transistors may be substituted forfield effect devices. The inputs to the differential pair amplifier arethe gates of MC1 and MC2. The sources of MC1 and MC2 are coupled incommon to a current source I_(tot). Current source I_(tot) is in turncoupled to a supply voltage V_(cc). Current source I_(tot) isconventional current source constructed as is known by those skilled inthe art.

The drains of MC1 and MC2 are coupled to current mirrors 7904 and 7906,respectively. Control voltage V_(c) is coupled to the gate of MC1 and avoltage reference is coupled to the gate of MC2. Voltage referenceV_(ref) is typically constructed as conventional voltage source known tothose skilled in the art. The currents present in the sources of MC1 andMC2 drive current mirrors 7904 and 7906, respectively. Current mirror7904 comprises transistors MC6 and MC3. Current mirror 7906 comprisestransistors MC4 and MC5. These current mirrors are constructedconventionally as is known by those skilled in the art. Output ofcurrent mirror 7904 and 7906 consists of the control signals iAtten andiSig.

FIG. 80 is a schematic of a VD1 control signal generation circuit.Control signal V_(c) is fed to the positive input of a differentialamplifier U2. Signal ended output of amplifier U2 is fed into the gateof transistor MC9. The source of MC9 is connected to the negative inputof U2. The source of MC9 is also coupled to a first terminal of aresistor R1. A second terminal of R1 is coupled to ground. The drain ofMC9 receives a current i_(c1) that is supplied by a drain of transistorMC7. The drain of MC7 is coupled to the gate of MC7. The source of MC7is coupled to a supply voltage V_(cc). The gate of MC7 is coupled to thegate of MC6. The source of MC6 is coupled to a supply voltage V_(cc).The drain of MC6 is coupled to a first terminal of a resistor R2. Thesecond terminal of resistor R2 is coupled to node 1001. The node formedby coupling MC6 to R2 supplies control signal VD1. Together transistorsMC7 and MC6 form a current mirror 8001.

Control current V_(c) sets up the control current i_(c1) throughamplifier U2, resistor R1 and transistor MC9. Current i_(c1) is mirroredthrough transistor MC7 and MC8 of the current mirror. The currentgenerated in the drain lead of MC6 creates a voltage across resistor R2as reference to the voltage present on node 7501. Thus, R1 and R2 aresized properly to control V_(ds) across M10, M4, M13 and M14. Forexample, VD1 can range from 100 mV to 600 mV. This condition correspondsto a V_(c)=05V at a minimum gain maximum input condition and a V_(c)=2.5V maximum gain minimum input signal condition.

In alternative embodiments, control voltage V_(c) may be subjected toconditioning by temperature compensation circuitry and linear in dBtransfer function compensation before being applied to the VD1generation circuit 7510.

The details of VGAs are disclosed in more detail in U.S. patentapplication Ser. No. 09/547,968 filed Apr. 12, 2000, (B600:36598)entitled “Large Gain Range, High Linearity, Low Noise MOS VGA” by AryaR. Behzad; based on U.S. Provisional Application No. 60/129,133 filedApr. 13, 1999 (B600:34440), the subject of which is incorporated in thisapplication in its entirety by reference.

What is claimed is:
 1. An integrated VCO circuit comprising: a VCOhaving a pair of driver transistors and a tank circuit; and a tuningcontrol circuit producing a sliding voltage window coupled to said tankcircuit for adjusting the frequency range of the VCO, said slidingvoltage window having a fixed voltage magnitude with a sliding centervoltage that is responsive to temperature so as to compensate fortemperature change in said pair of driver transistors.
 2. The integratedVCO circuit of claim 1, wherein said tuning control circuit comprises: aresistive divider circuit providing said sliding voltage window; a firstcomparator having a positive input coupled to a state signal and anegative input coupled to a first reference voltage of said slidingvoltage window; a second comparator having a positive input coupled tosaid state signal, and a negative input coupled to a second referencevoltage of said sliding voltage window; wherein outputs of said firstand second comparator control said tank circuit and thereby a resonantfrequency of said VCO; a bias transistor responsive to temperature andcoupled to said resistive divider circuit at a tap point between saidfirst and second reference voltages, wherein said bias transistoradjusts said sliding center voltage of said sliding voltage windowresponsive to temperature so as to compensate for a gate-to-sourcevoltage change with temperature in said pair of driver transistors. 3.An integrated VCO circuit comprising: a VCO having a pair of drivertransistors and a tank circuit; an adaptive bias circuit coupled to saiddriver transistors that maintains a nearly constant transconductance insaid pair of driver transistors in response to changing circuittemperature; and a tuning control circuit coupled to said tank circuitthat has a window of control voltages, said window having fixed voltagemagnitude with a sliding center voltage that is responsive totemperature; wherein maintaining a constant transconductance in saiddriver transistors assists in maintaining said sliding center voltage ofsaid window of control voltages.
 4. The integrated VCO circuit of claim3, wherein said VCO, said adaptive bias circuit, and said tuning controlcircuit are disposed on a common substrate.
 5. A method for minimizingnon-linear effects of temperature variation in an integrated VCO havinga tuning control circuit with a sliding voltage window, and a pair ofdriver transistors, the method comprising: sensing ambient circuittemperature; varying a gate-to-source voltage of each of the pair ofdriver transistors in response to the ambient circuit temperature tomaintain nearly constant transconductance in the pair of drivertransistors; and adjusting the sliding voltage window in response tosaid varying gate-to-source voltage of each of the pair of drivertransistors.
 6. An integrated VCO circuit comprising: a pair of drivertransistors; a tank circuit coupled to said pair of driver transistors;a tuning control circuit having a sliding voltage control window coupledto said tank circuit for setting a frequency tuning range; a means formaintaining a constant transconductance in said driver transistors overvarying temperature conditions; and a means for adjusting a centervoltage for said sliding voltage control window over varying temperatureconditions.
 7. The integrated VCO circuit of claim 6, wherein said apair of driver transistors, said tank circuit, said tuning controlcircuit, said means for maintaining a constant transconductance in saiddriver transistors, and said means for adjusting a center voltage forsaid sliding voltage control window are disposed on a common substrate.